a novel dual-band circularly polarized wide beam quadrifilar helix antenna

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圆极化TE_(11)输出模式旋向可在线切换的相对论磁控管模拟研究

旋圆极化 TE11 模式且模式纯度达到 99% 以上；当外加轴向磁场与微波轴向输出方向反向时，该器件的输出模式
即可在线切换为左旋圆极化 TE11 模式，而其他输出性能基本保持不变。
100 200 300 400 z/mm
Fig. 1 Schematic diagram of the longitudinal profile of the RM 图 1 RM 纵剖面结构
波导张角 θc2＝55°。该模型整体结构可分为四个部分，即 A 部分：用于电功率输入的同轴结构；B 部分：用于束波相互作用的谐振腔结构；C 部分：用于微波提取和模式转换的扇形波导结构；D 部分：用于微波输出的同轴结构。
Shi Difu， Qian Baoliang
（College of Advanced Interdisciplinary Studies, National University of Defense Technology, Changsha 410073, China）
Abstract： A relativistic magnetron with online switchable rotation direction of a circularly polarized TE11 output mode is proposed. In the device, the same cavity magnetron is adopted as the beam-wave interaction structure, the all-cavity extraction structure is adopted as the output structure, and the Helmholtz coils system is adopted as the magnetic system. In this paper, the output mode components of the device are theoretically analyzed by the mode excitation theory of all-cavity extraction structure, and the performance of the device is investigated by Particle in Cell simulation. Simulation results show that when the applied voltage is 770 kV and the applied magnetic field that has the same direction with the output microwave is 0.2 T, the device that operates at 5π/6 mode can output a right circularly polarized TE11 mode with the mode purity of more than 99%, the operating frequency of 2.35 GHz and the output power of 3.86 GW, corresponding to the power conversion efficiencyof 55.5%. When the direction of the applied magnetic field is reversed, the rotation direction of the right circularly polarized TE11 mode can be online switched to the left in the condition of keeping other performance of the device.

Hayati_et_al-2013-Microwave_and_Optical_Technology_Letters

2.88GHz.For the lower mode with C¼0.6pF and the higher mode with C¼5pF,the effects of varying/on CP perform-ance are given in Figures4and5,respectively.The simulation results suggest that an axial ratio of less than2dB can be found when/ranges between10and25 for the lower mode and between12and16 for the higher mode.3.RECONFIGURABLE DESIGN AND EXPERIMENTAL RESULTSAn antenna prototype with electrically switching was realized using a varactor diode(BB837,Siemens Semiconductor Group).For the dc bias(V0)used for controlling the varactor, its positive is connected to the feed line through a RF choke, which is composed of a high-impedance meandered microstrip line and a grounded capacitor of1nF,and the negative is directly linked to the RF ground plane,as shown in Figure1. Figure6exhibits the experimental results when V0is switched between two different values.From the measured results,it can be seen that the frequency with minimum axial ratio is1.83 GHz for the case of V0¼28V and it is2.96GHz for the case of V0¼6V.The CP bandwidths,determined by3dB axial ra-tio,are2.7and3.3%at the lower and higher CP operating fre-quencies,respectively.In addition,Figure6also demonstrates that a return loss of less than10dB is achieved within the two CP bandwidths.Therefore,the antenna can perform the dual-frequency operation with a frequency ratio of about1.6through switching.The radiation patterns at1.83and2.96GHz are measured and their results are plotted in Figure7.Broadside radiation with good CP performance is observed for each operating fre-quency,and the polarization in the plane of z>0is left-handed. The peak gain at1.83GHz is about3.3dBic and it is merely 0.2dB lower than that at2.96GHz.4.CONCLUSIONA design for circularly polarized annular slot antennas with switchable frequency has been presented.Only one diode is required in the reconﬁgurable design.With controlling the dc bias of the diode,the antenna can perform dual-frequency opera-tion with a high frequency ratio.Moreover,the antenna at the two operating frequencies has almost the same radiation pattern, polarization performance,and peak gain.REFERENCES1.Y.K.Jung and B.Lee,Dual-band circularly polarized microstripRFID reader antenna using metamaterial branch-line coupler,IEEE Trans Antennas Propag60(2012),786–791.2.Nasimuddin,Z.N.Chen,and X.Qing,Dual-band circularly-polar-ized S-shaped slotted patch antenna with a small frequency ratio, IEEE Trans Antennas Propag58(2010),2112–2115.3.J.Y.Sze,C.I.G.Hsu,and J.J.Jiao,CPW-fed circular slot antennawith slit back-patch for2.4/5GHz dual-band operation,Electron Lett42(2006),563–564.4.Y.L.Zhao,Y.C.Jiao,G.Zhao,Z.B.Weng,and F.S.Zhang,Anovel polarization reconﬁgurable ring-slot antenna with frequency agility,Microwave Opt Technol Lett51(2009),540–543.5.N.Jin,F.Yang,and Y.Rahmat-Samii,A novel patch antenna withswitchable slot(PASS):dual-frequency operation with reversed cir-cular polarizations,IEEE Trans Antennas Propag54(2006), 1031–1034.6.T.Y.Lee and J.S.Row,Frequency reconﬁgurable circularly polar-ized slot antennas with wide tuning range,Microwave Opt Technol Lett53(2011),1501–1505.V C2013Wiley Periodicals,Inc.DESIGN OF BROADBAND AND HIGH-EFFICIENCY CLASS-E AMPLIFIER WITH pHEMT USING A NOVEL LOW-PASS MICROSTRIP RESONATOR CELLMohsen Hayati1,2and Ali Lotfi11Electrical Engineering Department,Faculty of Engineering,Razi University,Tagh-E-Bostan,Kermanshah-67149,Iran; Corresponding author:mohsen_hayati@2Computational Intelligence Research Centre,Razi University,Tagh-E-Bostan,Kermanshah-67149,IranReceived31August2012ABSTRACT:In this article,a high-efﬁcient class-E ampliﬁer design with low voltage and broadband characteristics using a novel Front Coupled Tapered Compact Microstrip Resonant Cell is presented.The proposed micorstrip resonator is used as the harmonic control network in order to suppress higher order harmonics,which obtained the optimized impedance matching for the fundamental and harmonics.The class-E ampliﬁer is realized from0.7to1.8GHz,and obtained the power added efﬁciency of72.5–77.5%.The maximum value of Power added efﬁciency(PAE)is79.7%with11-dBm input power at1.5GHz. The designed class-E ampliﬁer using the proposed harmonic control network gained15.34%increment in PAE,and25.6%reduction in the circuit size in comparison with the conventional class-E ampliﬁer.The simulation and measurement results show the validity of the proposed design procedure of the broadband class-E ampliﬁer using a novel microstrip resonator cell.V C2013Wiley Periodicals,Inc.Microwave Opt Technol Lett55:1118–1118,2013;View this article online at .DOI10.1002/mop.27490Key words:switch mode;class-E amplifier;tapered cell;microstrip resonant cell;high efficiency;power added efficiency;zero voltage switching;zero voltage derivative switching1.INTRODUCTIONThe modern wireless communication systems need to consume the power supply.The main factor in reducing the consumption of the power supply is designing a low-voltage and high-efﬁ-ciency power ampliﬁer[1].The switch mode power ampliﬁer is an efﬁcient way for solving the efﬁciency problem.The class-E power ampliﬁer is a kind of the switch mode power ampliﬁer that the transistor acts as a switch.The class-E power ampliﬁer is tuned by a shunt capacitance.This type of the power ampliﬁer obtained100%drain efﬁciency theoretically[2].The class-E ampliﬁer’s response conditions are zero voltage switching (ZVS)and zero voltage derivative switching(ZVDS),which lead to zero power loss in the transistor.Therefore,a high-efﬁ-ciency power ampliﬁer is obtained[3].The shunt capacitance in the class-E power ampliﬁer has a main roll for achieving the class-E conditions[4].The power loss in the lower frequency can be neglected,but by increasing the operation frequency,the power dissipation is increased and the ideal operation of the class-E power ampliﬁer will be missed.The antiphase of the voltage and current wave-forms throughout the signal period,obtain the class-E power ampliﬁer with the maximum efﬁciency[5].This purpose can be achieved using a wave shaping network.The conventional class-E power ampliﬁer load resistance is very much lower than the transistor ON-resistance.This effect leads to efﬁciency degrada-tion and a narrowband load matching network[6].Furthermore, the transistor parasitic resistance for both the switch on-state and parasitic inductance leads to efﬁciency degradation in the radio frequency(RF)and microwave applications[7,8].The optimum operation of the class-E power ampliﬁer and the solution to the mentioned drawbacks can be obtained using two main methods:namely active device selection and circuit conﬁguration[9].The class-E ampliﬁer has various conﬁgura-tions such as the cascade[10]and push–pull[11].The cascade class-E conﬁgurations can double the maximum permissible drain voltage,and the push–pull class-E conﬁguration increases the output power and decrease the harmonic distortion with high efﬁciency.A new topology for the class-E ampliﬁer is proposed as an inverse class-E ampliﬁer,which has inductive reactance [12].The inverse class-E ampliﬁer has higher load resistance and lower peak switch voltage in comparison with the class-E ampliﬁer.Also,because of the abruption of the device output inductances,the value of the inductance in the load network is decreased.However,the inverse class-E ampliﬁer can be used only for the small to medium power applications.Therefore,to solve this drawback,the power combining methods have been used[13].Although,this method results in obtaining the inverse class-E ampliﬁer for higher power application,but the circuit conﬁguration and the design procedure are complicated with the circuit size increment because of using two power ampliﬁer circuits.The class-E power ampliﬁer is a high-efﬁciency power am-pliﬁer for the microwave application,which is implemented using the transmission line as the harmonic control network at the output of the ampliﬁer circuit[14].Furthermore,instead of the RF choke(RFC)a section of the transmission line is used.The transmission line has been used in the class-E power ampliﬁer using LDMOS[15],GaN HEMT[16–19],SiC MES-FET[20],and LDMOSFET[21]as the harmonic control net-work increasingly,because of the simplicity of its structure and high rejection of harmonics.Therefore,the class-E ampliﬁer conﬁguration and operation are the best candidates for the design of the ampliﬁer for the modern microwave communica-tion systems[22,23].Consequently,designing of the load network as the harmonic control network for suppression of harmonics in order to obtain a high-efﬁciency power ampliﬁer is the main challenge of the switch mode power ampliﬁers.The designing of the class-E power ampliﬁers using various microstrip structures has been proposed such as a defected ground structure[24],an asymmet-rical spur-line[25],and composite right/left-handed transmission lines[26].The narrowband load network and low efﬁciency remain as the main challenge to the class-E power ampliﬁer using the conventional microstrip transmission line[27].A compact microstrip resonant cell(CMRC)is a one-dimen-sional photonic band gap incorporating the microstrip transmis-sion line,which is,ﬁrst,proposed in[28].The CMRC structure exhibits high rejection of the harmonics with the compact circuit size in comparison with the conventional micorstrip transmission lines.Therefore,it is used for the linearization and efﬁciency in-crement of the microwave power ampliﬁers[29,30].The appli-cation of the conventional CMRC is limited to obtain a high-ef-ﬁciency switch mode ampliﬁer,as a result of the high insertion loss in the passband and restricted stopband.The front coupled tapered CMRC(FCTCMRC)is proposed in[31]for the implan-tation of a low-passﬁlter with high and wide rejection in the stopband with the compact circuit size in comparison with the conventional CMRC.Therefore,it can be widely used for designing the high-efﬁciency and broadband switch mode power ampliﬁer because of high and wide suppression of harmonics.In this article,the harmonic suppression of the class-E ampli-ﬁer using a novel FCTCMRC as the harmonic controller net-work is explored.A class-E ampliﬁer with higher efﬁciency at a wider bandwidth in comparison with the conventional ampliﬁers is achieved.The proposed class-E power ampliﬁer is designed and simulated for a frequency of1.5GHz using the micorstrip resonator structure.The measurement results of the proposed power ampliﬁer validate our design procedure and simulation results.2.CLASS-E AMPLIFIER FUNDAMENTAL AND DESIGN THEORY2.1.Class-E Ampliﬁer OperationThe basic circuit conﬁguration of the class-E ampliﬁer and switch waveforms are shown in Figures1(a)and1(b),respec-tively.The class-E ampliﬁer consists of the switch device,shunt capacitance,series-tuned load network L-C,and an ideal RFC. The switch-on duty ratio is assumed to be50%in designing the class-E ampliﬁer.This value of the duty ratio leads to optimum operation of the class-E ampliﬁer for obtaining high efﬁciency [32].For an ideal class-E operation,three requirements for the drain voltage and current should be met[2]:1.The rise of the voltage across the transistor at turn-offshould be delayed until the transistor is off.2.The drain voltage should be brought back to zero at thetime of the transistor turn-on.3.The slope of the drain voltage should be zero at the timeof the transistor turn-on.Therefore,the class-E power ampliﬁer is constructed based on two conditions as ZVS and ZVDS.These conditions are as follows:v s hðÞjh¼p¼0;(1)dv s hðÞd hh¼p¼0;(2)where v s(y)is the switch voltage,and y¼x t.The quality fac-tor of the output series resonant circuit is assumed inﬁnite. Therefore,the output current is sinusoidal asi oðhÞ¼I m sinðhþuÞ:(3)In the time interval0y<p,the switch device is in the on-state,therefore,using Kirchhoff’s current law at the switch,we havei sðhÞ¼I dc1þa sin hþuðÞðÞ:(4)This is the currentﬂow through the shunt capacitance in the switch-off state.Therefore,the voltage across the switchisFigure1(a)The basic circuit of the class-E ampliﬁer.(b)The class-E switch voltage and current waveformv sðtÞ¼1C sZ ti sðt0Þdt0¼I dcx C s1þa cos x tþuðÞÀcos uðÞðÞ:(5)Applying the class-E ZVS and ZVDS conditions to Eqs.(4)and (5),the value of a and u can be obtained asa¼ﬃﬃﬃﬃﬃﬃﬃﬃﬃﬃﬃﬃﬃ1þp24r;(6)u¼ÀtanÀ12p8>:9>;:(7)The drain voltage waveform is shaped by the harmonics so that the drain voltage and the slope of the drain voltage is zero when the transistor is in the on-state.The reactance for all harmonics is negative and comparable in magnitude to the fundamental fre-quency load resistance.The ideal class-E ampliﬁer requirements are difﬁcult to meet.So,we often only tuned the second and third harmonics to get the suboptimum class-E power ampliﬁer solution.The analysis is performed considering just the output network behavior,thus neglecting input signal required to oper-ate the active device as an ideal switch.The optimal fundamental load by the Fourier-series expan-sion analysis in[7]used for achieving the perfect class-E opera-tion can be determined asZ E;f0¼0:28x C Pe49 :(8)This impedance is inductive.On the other hand,for the ideal operation of the class-E power ampliﬁer the impedances at the higher order harmonics are inﬁniteZ E;fn¼1;for n!2:(9)From(8)the nominal class-E ampliﬁer shunt capacitance C is deﬁned byC¼0:1836x0R:(10)In order to achieve the maximum operation frequency of the class-E ampliﬁer,the device output capacitance should be equal to Eq.(10).The matching network for the class-E power ampli-ﬁer using a low-pass Chebyshev-form impedance transformer is proposed in[7].Therefore,the synthesis of the load network is done using a short circuit,and open circuit stubs instead of lumped capacitors in the load network for unwanted harmonics.2.2.Design of a Class-E Ampliﬁer Using a pHEMTAchieving the optimum load is the main factor to obtain high efﬁciency when designing the class-E power ampliﬁer.On the other hand,the optimum load is varied with the operating fre-quency as in Eq.(8).Therefore,designing of the load network, which can operate in the wide frequency range,is needed for designing the class-E power ampliﬁer with the optimum condi-tions.The maximum operation frequency of the class-E power ampliﬁer is restricted by the shunt capacitance.The shunt capac-itance consists of the transistor output capacitance and the exter-nal capacitance.Thus,the optimum operating frequency of the class-E power ampliﬁer is achieved by selecting a transistor with lower output capacitance.On the other hand,the power loss is caused by ON-resistance of the transistor[33].Therefore, the active device with lower ON-resistance is preferred for designing the high-efﬁciency class-E power ampliﬁer.We selected an ATF-34143pHEMT because of its lower ON-resist-ance and lower shunt parasitic capacitance,which provides lower power dissipation and optimum operation frequency using external capacitance,respectively.The circuit topology of the conventional class-E ampliﬁer is shown in Figure2(a).It is designed using the design procedure,which is presented in[2, 3].The value of elements for an ideal class-E power ampliﬁer is tabulated in Table1.In the design of the class-E power ampli-ﬁer,it is assumed that the value of the DC-feed is inﬁnitive,but in real implementation this value isﬁnite,and we used the half wavelength microstrip transmission line for the DC-feed.In the conventional class-E ampliﬁer,using lumped elements, the second harmonic is located within the pass band.Therefore, the bandwidth is limited to one octave.In order to solve this drawback,one way is designing a multiple matching network for various bands and using switching element.This way leads to complexity of the ampliﬁer circuit and degradation of the efﬁciency.The use of the micorstrip transmission line is a low-cost and simple way for designing the class-E ampliﬁer with wide band and high-efﬁciency characteristics.We used the design proce-dure in Section2.1and designed the matching network for the ampliﬁer as shown in Figure2(b).The values of the transmis-sion lines dimensions are given in Table2.The class-E ampli-ﬁer is designed on RT/Duroid5880,a substrate with dielectric constant of2.2,height of15l l,and loss tangent of0.0009.Figure2Idealized class-E power ampliﬁer:(a)lumped elements and(b)transmission lineTABLE1Element Design for the Nominal Class-E AmplifierC i1(pF)C i2(pF)C o1(pF)C o2(pF)C e(pF)C g1(pF)C g2(pF)C d1(pF)C d2(pF)L i1(nH)L o1(nH)L o2(nH) Theoretical10010010010 4.2221000.50.2312 4.7 3.33.FRONT COUPLED TAPERED CMRC CHARACTERISTICSA novel FCTCMRC is proposed in [31],for the ﬁrst time,which is used to synthesize a low-pass ﬁlter with high and wide rejec-tion in the stopband.This microstrip structure exhibits bandstop characteristics and slow wave effects,which are used in the stopband extension and the circuit size reduction,respectively.The schematic and equivalent circuit of the resonator is shown in Figures 3(a)and 3(b),respectively.The proposed FCTCMRC has symmetrical topology.Therefore,the even–odd mode [34]can be used to simplify the analysis as shown in Figures 3(c)and 3(d).Consequently,theresonant condition for the odd-mode in Figure 3(c)is obtained by equating the input admittance Y o in of the proposed resonator to zero yields:Z 112x C 1ÀZ 1tan h 1 ÀZ 2tan h 2Z 1þtan h 12x C 1¼0:(11)Using the similar procedure,the even-mode resonant frequencies areobtained by equating the even admittance Y e in to zero as follows:Z 2tan h 1þZ 1tan h 2¼0:(12)The transmission zeros of the equivalent circuit for the proposed FCTCMRC,which is shown in Figure 3(a),is obtained whenY o in ¼Y ein asZ 2sin 2h 2þZ 1sin 2h 1¼cos 2h 1x C 1:(13)Therefore,the resonator characteristics for tuning transmission zeroes in the stopband can be achieved by the length and width of the tapered cells as shown in Figures 4(a)and 4(b).The pro-posed structure is optimized by an EM-simulator (ADS).The obtained dimensions are as follows:L t1¼2:58;L 2¼1:94;L 3¼2:7;W t1¼2:71;W t2¼5:6;W 1¼0:1;W 2¼0:56;L 3¼0:75;L f ¼2:36;W f ¼0:25all are in millimeter ðmm Þ:TABLE 2The Value of the Conventional Transmission Line for the class-E AmplifierTL 1TL 2TL b1TL 3TL 4TL 5TL b2Width (mm) 4.730.940.620.71 1.24 4.210.72Length (mm) 6.319.7262.3137.2318.4264.3Figure 3(a)Schematic of the proposed resonator.(b)Equivalent cir-cuit.(c)Odd-mode.(d)EvenmodeFigure 4(a)Changing of the transmission zeros with the width of tapered cell W t1.(b)Changing of the transmission zeros with the length of tapered cell L t .(c)Simulation and measurement results of the proposed harmonic control network.(d)Simulation input impedance of the FCTCMRCThe proposed FCTCMRC is fabricated,and the measurement is performed using an Agilent N5230A Network Analyzer.The simulation and measurement results of the proposed FCTCMRC are shown in Figure 4(c).As it is shown,it has an attenuation level À43and À33.1dB at 3.0and 4.5GHz,respectively.Therefore,the high suppression for the second and third har-monics is obtained.The insertion loss from DC to 2.39GHz is lower than À0.1dB.The simulation of the input impedance of the proposed CMRC for the fundamental and harmonics is shown in Figure 4(d).As it is observed,the harmonic impedan-ces are relatively open in comparison with the fundamental im-pedance.Consequently,it can be used as the matching network with high performance and low circuit complexity.4.CIRCUIT DESIGN AND IMPLEMENTATIONThe highly efﬁcient and compact size class-E ampliﬁer is designed and implemented for a 1.5-GHz band using an ATF-34143pHEMT.The proposed circuit is simulated using an Agi-lent’s Advanced Design System (ADS),and fabricated on an RT/Duroid 5880substrate.The active device is biased at V d ¼3V and V g ¼À0.7V.The FCTCMRC is used as the harmonic control network (HCN)at the output of the active device.The proposed HCN absorbed the parasitic reactance and capacitance of the active device.Therefore,it does not need to any lumped elements in series or parallel with the transistor to compensate the parasitic elements.The circuit schematic diagram of the designed class-E ampliﬁer is shown in Figure 5(a).Moreover,the photograph of the fabricated circuit is shown in Figure 5(b).The RFC is realized using the microstrip transmission line (TLb2)with the quarter wavelength at a frequency of 1.5GHz.The input matching elements consist of two series and parallel open stubs.The dimensions of the tapered cells and transmission lines in the HCN are tuned in order to optimize harmonic termi-nation in the implemented ampliﬁer circuit.The design and implementation of the output matching networks using the FCTCMRC as low-pass topology has been done from 0.7to 1.8GHz.The voltage and current waveforms of the designed class-E ampliﬁer are shown in Figure 5(c).The switch is open for the time interval,0.2–0.4ns and the current through it is near zero.The switch is closed during the time interval 0.6–0.8ns,and the voltage across it is near to zero.The class-E ZVS and ZVDS conditions in the switch turn-off state are obtained.Therefore,the high-efﬁciency class-E ampliﬁer is achieved.The input signal is generated using an Agilent E4433B signal generator,and the measurement is done by an E4440A PSA se-ries spectrum analyzer.The simulated and measured output power and gain for P in ¼11dBm (input power)are shown in Figure 6(a).The maximum output power at 1.5GHz with P in ¼11dBm is 25.3dBm,and the related gain is 14.3dB.The con-ventional class-E ampliﬁer without CMRC has an output power of 18.5dBm and gain of 7.5dB.The class-E ampliﬁer using CMRC has 36.7%output power improvement in comparison with the one without CMRC.The simulation and measurement results for the PAE at P in ¼11dBm (input power)is shown as a function of the operating frequency in Figure 6(b).The highest value of PAE at a fre-quency of 1.5GHz was 79.7%.The value of the PAE is 69.1%for the conventional class-E ampliﬁer without CMRC.There-fore,the proposed class-E ampliﬁer using the novel CMRC has 15.34%PAE improvement in comparison with the one without CMRC.The output power of the conventional class-E ampliﬁer is decreased as the operating frequency is increased.As shown in Figure 6(a),this decrement is considerable when the operating frequency is more than 1.2GHz.Therefore,the conventional class-E ampliﬁer has a drawback for the broadband applications.The designed class-E ampliﬁer has 25.6%circuit size reduction in comparison with the conventional class-E ampliﬁer.5.CONCLUSIONThe class-E ampliﬁer with high efﬁciency and broadband char-acteristics has been designed and implemented.A novel and simple load-matching technique for the low-voltage microwave class-E ampliﬁer using a front-coupled taperedcompactFigure 5The pHEMT class-E ampliﬁer.(a)Circuit conﬁguration.(b)A photograph of fabricated ampliﬁer.(c)Simulated switch voltage and current waveforms.[Color ﬁgure can be viewed in the online issue,which is available at ]microcstrip resonant cell has been presented.The proposed am-pliﬁer achieved an output power of 25.3dBm,a power added efﬁciency of 79.7%,and a gain of 7.5dB at input power of 11dBm.It has high-efﬁciency performance over a signiﬁcant band-width form 0.7to 1.8GHz (88%).The proposed compact micro-strip resonant cell as the harmonic control network exhibited 15.34%improvement in PAE and 25.6%reduction in the circuit size in comparison with the conventional class-E ampliﬁer.The extremely low insertion loss at the fundamental frequency and size reduction characteristics can be used in the design of the class-E ampliﬁer with higher output power and smaller size,which are required in the 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用于Ka波段卫星通信的双频段圆极化无源相控阵天线

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低RCS宽带磁电偶极子贴片天线设计

低RCS宽带磁电偶极子贴片天线设计张晨;曹祥玉;高军;李思佳;黄河【摘要】该文设计了一种低雷达散射截面(RCS)的宽带磁电偶极子贴片天线，其中印刷在介质板上的金属贴片为电偶极子，3个金属过孔连接辐射贴片与金属地板构成磁偶极子。

整个天线采用“T”型渐变馈电结构同时激励电偶极子与磁偶极子，天线的频带范围为7.81~13.65 GHz，覆盖了整个X波段。

实测和仿真结果表明，通过在磁电偶极子贴片天线底面采用开槽技术并优化开槽的形状、大小、位置等变量，在天线工作频带范围内实现了RCS的减缩，最大缩减量达到了17.9 dB，同时天线保持了增益稳定不变，E面、H面方向图一致的特性。

%A low Radar Cross Section (RCS) and broadband Magneto-Electric (ME) dipole patch antenna from 7.81 GHz to 13.65 GHz covering the whole X band is designed and fabricated. Metal patches printed on the substrate form the electric dipoles, three metallic vias connected to the radiation patches and the metal ground account for the magnetic dipole radiation. The whole antenna is connected with a T-shaped feed structure which excites electric and magnetic dipoles simultaneously. Numericaland experimental results incident that the RCS of the ME dipole patch antenna can be reduced inthe whole bandwidth which the largest value is up to 17.9 dB by cutting slots on the ground and optimizing the size, shape, position of the slots. Also, the antenna shows advanced performances such as stable gain and almost consistent pattern in E and H plane.【期刊名称】《电子与信息学报》【年(卷),期】2016(038)004【总页数】5页(P1012-1016)【关键词】磁电偶极子天线;宽频带;开槽技术;低RCS;一致性【作者】张晨;曹祥玉;高军;李思佳;黄河【作者单位】空军工程大学信息与导航学院西安 710077;空军工程大学信息与导航学院西安 710077;空军工程大学信息与导航学院西安 710077;空军工程大学信息与导航学院西安 710077;西安通信学院西安 710106【正文语种】中文【中图分类】TN821 引言微带贴片天线以其低剖面、易共形等优点在战场通信、监视及其它作战平台上得到了广泛应用，但由于带宽窄，不能用于宽频天线系统，且E面、H面方向图差异较大，不易于组成天线阵[1,2]。

应用于GNSS终端的宽带扁平状双频圆极化微带天线

通信技术终端的宽带扁平状双频圆极化微带天线，李晓鹏1，李成钢2，蔡惠萍广州中海达卫星导航技术股份有限公司，广东广州511400；2.广州市中海达测绘仪器有限公司，广东终端的宽带扁平状双频圆极化微带天线。

通过在天线辐射单元基板外围设置短路加载振子，不仅可以降低天线谐振频率实现天线的小型化设计，而且可有效提升天线辐射增益和工作带宽。

双频天线单元分别采用四馈点馈电方式，使得天线拥有稳定且可靠的相位中心偏差值和良好的圆极化特性。

针对该天线设计模型，使用仿真软件对天线进行仿真，可以得出天线在高频段（1.1661.607 GHz）内的各项指标表现较优。

通过制作实物样机进行实际测量，结果表明该天线在上述高低频双1.9 dBi，对应高低频段内中心频点的辐射增益大于180°。

因此，该天线能够较好满足目前GNSS短路加载；小型化；宽频段；相位中心；高增益Wide-Band Flat Dual-Band Circularly Polarized Microstrip Antenna for GNSSApplications，LI Chenggang2，CAI Huiping.Hi-Target Navigation Tech Co.，Ltd.，Guangzhou（1）（2）r为相对天线馈电点的位置可由公式粗略估算，然后通。

给出输入电阻与馈点（3）为天线馈点距离中心点位置；h0为平均电场强度；P为辐图1 天线结构俯视图2 仿真与实测分析本文采用专业电磁仿真软件对天线模型进行仿真优化，并使用优化后的结构参数将天线制作实物样品，如图图2 天线实物图图为实测的驻波比随频率变化曲线，在整个测试频段1～1.8 GHz内，天线的驻波比均小于说明该天线具有良好的阻抗匹配特性。

图线轴比随频率变化曲线，在整个测试频段，天线的轴，说明天线的圆极化性能好。

图3 天线实测驻波比随频率变化曲线图4 天线仿真与实测的轴比随频率变化曲线图5为天线仿真与实测的增益随频率变化的曲 2020年11月25日第37卷第22期Telecom Power TechnologyＮｏｖ．　２５，２０２０，Ｖｏｌ．　３７　Ｎｏ．　２２　林飞，等：应用于GNSS终端的宽带扁平状双频圆极化微带天线线。

宽频带圆极化交叉偶极子天线设计

宽频带圆极化交叉偶极子天线设计卢红;张文梅;韩丽萍;薛哲;陈新伟【摘要】本文设计了一种宽频带圆极化的交叉偶极子天线,天线由两组正交放置的加载 L型分支线的矩形偶极子、同轴馈线和矩形反射板构成.两个 3/4 圆环插入到两组正交偶极子天线之间,实现圆极化特性.通过在矩形交叉偶极子上增加两条分支线和在边缘切角有效拓宽了天线的阻抗带宽和轴比带宽.仿真结果表明:设计天线的相对阻抗带宽达到 1.5~5.2 GHz(110.4%),3-dB轴比带宽达到 1.67~3.77GHz(77%),增益最高可达 7 .2 dBi,故设计的天线可被用于无线能量收集系统.%In this paper,a broadband circularly polarized cross dipole antenna is designed.The pro-posed antenna consists of two sets of rectangular dipoles which are orthometric and loaded with L-type branchinglines,coaxial feed line and rectangular reflector plane.Two 3/4 rings are inserted into the two orthogonal dipole antenna to realize the circular polarization characteristics.The impedance band-width and the axial ratio bandwidth of proposed antenna are effectively broadened by adding two branch lines in each group of rectangular cross dipole antenna and cutting a triangle at the edge of the dipole. The simulated results show that the antenna has an impedance bandwidth of 1.5 to 5.2 GHz (110.4% ) ,and axial ratio bandwidth can be up to 1.67 to 3.77 GHz (77%),the maximum gain can be up to 7.2 dBi.The proposed antenna can be used in wireless energy harvesting system.【期刊名称】《测试技术学报》【年(卷),期】2018(032)004【总页数】5页(P344-348)【关键词】宽频带;圆极化;交叉偶极子;天线【作者】卢红;张文梅;韩丽萍;薛哲;陈新伟【作者单位】山西大学物理电子工程学院,山西太原 030006;山西大学物理电子工程学院,山西太原 030006;山西大学物理电子工程学院,山西太原 030006;山西大学物理电子工程学院,山西太原 030006;山西大学物理电子工程学院,山西太原030006【正文语种】中文【中图分类】TN828.6近年来，无线能量收集系统已经引起了国内外的广泛关注，其中天线是整个系统的关键部分之一，很多文献对接收天线进行了研究[1-5]. 文献[1]设计的接收天线实现了5.8 GHz的能量收集，但空间环境中电磁波信号的功率密度比较低，并且分布在多个频带. 文献[3]设计了可以在0.55, 0.75, 0.9, 1.85, 2.15和2.45 GHz六个频带工作的天线，但天线结构比较复杂. 文献[4]利用两组扇形交叉偶极子实现了1.7～3.0 GHz的宽频带双极化天线. 然而，空间中电磁波的极化方向是随时改变、不可预测的，线极化和双极化天线都会造成比较大的极化失配损耗，因此，宽频带圆极化天线在能量收集方面更受青睐. 文献[6]采用两组末端比较宽的矩形交叉偶极子，实现了2.3～2.9 GHz宽频带内的圆极化特性. 文献[7]利用印刷在介质基板上层的4个开口谐振环作为条形交叉偶极子的寄生单元，实现了较宽圆极化带宽和阻抗带宽. 文献[8]中,印刷在同一介质基板上的两个倒钩形偶极子和两个蝴蝶形交叉偶极子形成了双宽带圆极化天线. 文献[9]通过使用不对称的蝴蝶结交叉偶极子使圆极化带宽提高到51%，阻抗带宽提高到57%. 文献[10]采用了阶梯式矩形交叉偶极子结构和不规则的地，使轴比带宽和阻抗带宽分别扩大到55.1%和66.9%. 文献[11]采用一个寄生单元和简单的蝴蝶结偶极子天线将圆极化带宽和阻抗带宽提高到58.6%和68.9%. 这些文献虽然都实现了宽频带的圆极化性能，但所实现的频段没有同时覆盖GSM、LTE、WLAN等频段，为了能量收集系统能够尽可能多地收集空间中的能量，需要设计阻抗带宽和轴比带宽更宽的天线.本文介绍了一种可以用于能量收集的宽频带圆极化交叉偶极子天线. 通过在两组交叉偶极子之间插入3/4圆环来实现圆极化特性，在每组矩形交叉偶极子天线上增加两条分支线，同时在偶极子的边缘切角，进而有效拓宽天线的阻抗带宽和轴比带宽.1 天线设计本文设计的宽频带圆极化天线如图 1 所示. 天线由两组垂直交叉放置的带L形分支线的矩形偶极子、两个3/4圆环、一个反射板、同轴馈线构成. 两组偶极子分别印刷在相对介电常数为2.2，损耗角为0.000 9，厚度为31 mil，大小为80×80 mm2的RT/Duroid5880介质基板的两面. 其中，两组偶极子由4个矩形贴片加载两对L型分支线和在边缘切角变化而来. 印刷在介质基板中心正反面的两个3/4的圆环分别连接两组偶极子的一端，形成90°相位差. 整个天线的上下层通过50 Ω同轴线馈电，上层与同轴线的内芯相连，下层与同轴线的外层相连，天线放置在距离反射板H处.图 1 天线结构图Fig.1 Configuration of antenna天线各部分参数如表 1 所示.表 1 天线的各部分尺寸Tab.1 The values of the antenna参数L3W3L4W4L1W1L21数值/mm808090902320.25.5参数L22W2RinWrdb(a)H 数值/mm2.31.23.41.10.5735设计的天线由两组矩形偶极子天线变化而来，将只有两组矩形偶极子的天线作为天线1；在矩形贴片的偶极子上增加一组L型分支线的天线作为天线2；带有L型分支线的偶极子天线2的边缘切角后的天线作为天线3(本文提出的天线). 图 2 给出了天线1，天线2和天线3的回波损耗和轴比图. 从图2(a)中可以看出，在天线1的基础上增加L型分支线和切角后，在高频段引入了新的谐振点，有效地拓宽了天线的阻抗带宽；从图2(b)可以看出，增加分支线后整个频带内的轴比减小，两组偶极子的边缘切角使得两个圆极化谐振点远离，且两个谐振点之间频带内的轴比都小于-3 dB，有效地拓宽了轴比带宽.图 2 各参考天线回波损耗S11和轴比图Fig.2Simulated return loss and aixal ratio of the reference antennas2 参数分析图 1 中提到的宽带圆极化交叉偶极子中，所加载的L型分支线由两部分构成，其中长度l21对天线的阻抗带宽和轴比带宽有明显的影响，长度l22的变化对天线性能影响不大，在偶极子天线边缘切去的三角形中，边长b的长度对天线的性能有明显影响，在分析某一参数对天线性能影响时，其它参数均保持不变. 图 3 给出了L形分支线中l21的长度对天线回波损耗S11和轴比带宽的影响. 图 4 给出了切去三角形的边长b对天线S11和轴比带宽的影响.图 3 l21对天线性能的影响Fig.3The effect of parameter l21 on antenna performance图 4 b对天线性能的影响Fig.4The effect of parameter b on antenna performance从图 3 可以看出，随着l21的减小，阻抗带宽变宽，通带内的匹配特性变好，两个圆极化谐振点远离，且两个谐振点之间的频带内的轴比逐渐减小. 从图4中可以看出，切角b的改变对天线S11基本没有影响，但对轴比带宽影响较大. 随着b 的减小，高频段的谐振点保持不变，低频段的谐振点向低频移动，有效的拓宽了天线的轴比带宽，所以b选择为7 mm.3 优化结果根据图 1 所设计的宽频带圆极化天线，在软件HFSS中建立模型并进行电磁全波仿真. 通过参数分析后，得出了如表 1 所示的最优参数. 图 5～图 7 给出了最优参数下宽频带圆极化天线的仿真结果.图 5 天线的S11和轴比曲线Fig.5 S11 and axial ratio (AR) of the antennas 图 6 天线的增益Fig.6 Realized gain of the antennae图 7 天线的辐射方向图Fig.7 Radiation patterns of the antenna从图 5 可以看出，设计的天线在1.5～5.2 GHz频带内，该天线的S11都小于-10 dB，在1.67～3.77 GHz 频带内，该天线的轴比都小于3 dB. 从图 6 看出，1.5～3.5 GHz增益都可大于5 dBi，随着频率的升高，天线的增益有所降低. 图 7 给出了1.8，2.6，3.5 GHz处的辐射方向图，从图中可以看出右旋圆极化大于左旋圆极化，因此，所设计天线为右旋圆极化天线.4 结论本文设计了一种可用于能量收集的宽频带圆极化天线. 通过在每组矩形交叉偶极子天线上增加两条分支线和在偶极子边缘切掉一个三角形来有效拓宽天线的阻抗带宽和轴比带宽. 仿真结果表明：设计的天线阻抗带宽为1.5～5.2 GHz(110.4%), 轴比带宽为1.67～3.77 GHz(77%), 包含了GSM，LTE和WLAN所有的频带，轴比带宽范围内最高增益可达7.2 dBi.参考文献：【相关文献】[1] Mcspadden J O, Fan L, Chang K. Design and experiments of a high-conversion-efficiency 5.8 GHz rectenna[J]. Microwave Theory & Techniques IEEE Transactions on, 199 8, 46(12)： 2053-2060.[2] Kuhn V, Lahuec C, Seguin F, et al. A multi-band stacked RF energy harvester with RF-to-DC efficiency up to 84%[J]. IEEE Transactions on Microwave Theory & Techniques, 2015, 6 3(5)： 1768-1778.[3] Song C, Huang Y, Carter P, et al. A novel six-band dual CP rectenna using improved impedance matching technique for ambient RF en ergy harvesting[J]. IEEE Transactions on Antennas & Propagation, 2016, 64(7)： 3160-3171.[4] Suh Y H, Chang K. A high-efficiency dual-frequency rectenna for 2.45 and 5.8 GHz wireless power transmission[J]. Microwave Theor y & Techniques IEEE Transactions on, 2002, 50(7)： 1784-1789.[5] Nakano H, Kikkawa K, Kondo N, et al. Low-Profile Equiangular Spiral Antenna Backed by an EBG Reflector[J]. Transactions on Antenn as & Propagation, 2009, 57(5)： 1309-1318.[6] He Y, He W, Hang W. A wideband circularly polarized cross-dipole antenna[J]. IEEE Antennas & Wireless Propagation Letters, 2014, 13(1)： 67-70. [7]Baik J W, Lee T H, Pyo S, et al. Broadband circularly polarized crossed dipole with parasitic loop resonators and its arrays[J]. IEEE Transactions on Antennas & Propagation, 2011, 59( 1)： 80-88.[8] Tran H H, Park I. A dual-wideband circularly polarized antenna using an artificial magnetic conductor[J]. IEEE Ante nnas & Wireless Propagation Letters, 2016, 15(7)： 950-953.[9] Tran H H, Park I. Wideband circularly polarized cavity-backed asymmetric crossed bowtie dipole antenna[J]. IEEE Antennas & Wireless Propagati on Letters, 2016, 15(3)： 358-361.[10] Yang W, Pan Y, Zheng S, et al. A low profile wideband circularly polarized crossed-dipole antenna[J]. IEEE Antennas & Wireless Propagation Letters, 2017, 16(5)： 2126-2129.[11] Tran H H, Park I, Nguyen T K. Circularly polarized bandwidth-enhanced crossed dipole antenna with a simple single parasitic element[J]. IEEE Antennas & Wireless Propagation Letters, 2017, 16(3)： 1776-1779.。

高效率双圆极化天线阵

面阵进行电磁仿真验证，天线阵由2×4个双极化单元组成，如图2所示，4单元左右旋圆极化线阵各两根，两种极化线阵相间位于同一平面排列，其中每一圆极化阵中的水平极化辐射缝相同，而垂直极化辐射缝偏置相反，如图2(b)所示。

1(a)三维立体图 1(b)俯视图图1 圆极化辐射单元结构示意图为了减轻天线重量、满足两种极化阵间距限制条件，相邻极化线阵共用波导壁厚，水平极化缝隙切割同一波导壁，保证辐射缝隙的谐振长度。

2 仿真结果验证设计思路选择工作于X 频段，使用HFSS 仿真软件，先就单圆极化线阵进行优化设计，另一极化与其相同，仅纵向长缝偏置位置差异，单圆极化优化仿真完成后，两种极化0 引言圆极化天线在通信、遥测、导航、目标探测和SAR 中都有广泛的应用。

应对系统不同需求，圆极化天线发展出多种形式，如：螺旋、微带天线、十字交叉振子、喇叭、缝隙波导和开口波导等。

在圆极化阵列无线中，缝隙波导天线常采用集中式馈电结构，波导馈电网络和缝隙天线采用一体化设计和加工，具有结构简单，加工容易，结构强度高，高效率和功率容量大等优点。

特别是在Ku、Ka 及更高频段，发展出多种波导馈电天线阵[1-2]。

缝隙波导在行波中产生两种极化分量且有90°相差，就能得到圆极化。

本文给出一种左右旋双圆极化缝隙波导天线阵设计方法。

单元结构见图1，在一段对称脊波的窄边上开两个对称的倾斜缝，在凹陷的宽边上开纵向偏置缝，通过脊高度的合理选择，凹陷部分的高度约为工作波长的四分之一，即可得到圆极化。

对于与之正交的圆极化，凹陷内部纵向缝偏置波导壁的另一侧即可以实现。

由于圆极化线阵建立在压缩宽度的脊波导基础上，因此两种圆极化线阵可以相间排列，如图2所示。

1 缝隙波导阵结构the RHCP and LHCP is the longitudinal slot offset. The dual CP antenna array obtains high efficiency and simple structure. A simple planar array is simulated to verify the design.Keywords: Circularly polarization;waveguide;antenna array图4 带内轴比频率响应5(a)8.1GHz5(b)8.25GHz5(c)8.4GHz图5 左旋圆极化方向图6(a)8.1GHz6(b)8.25GHz,6(c)8.4GHz图6 右旋圆极化方向图（下转第3.1 明确主接线设计要求和具体情况在进行220KV 变电站电气一次主接线设计的时候，首先必须要明确变电站电气一次主接线的具体要求和变电站的具体情况，通过实地考察获得一手资料，最终本着“灵活性、可靠性、经济性”的原则来展开一次主接线设计工作，最终获得最佳的主接线方案。

单馈双频GPS微带天线

小型化双频段GPS微带天线*彭祥飞1，钟顺时1，许赛卿2,1，武强1（1.上海大学通信及信息工程学院，上海 200072；2. 浙江正原电气股份有限公司浙江嘉兴 314003）摘要：最近为了满足GPS定位准确性和可靠性的需要，要求天线在GPS两个频率上实现圆极化。

本文介绍一种通过单个探针馈电的双层正方形切角的微带贴片天线，采用不同介电常数的微波陶瓷基片。

及常规的双频圆极化天线相比，天线尺寸减小了且没有在两层贴片间引入空气层，结构紧凑，便于加工。

文中给出天线的详细设计及实验结果，并进行了讨论，实测结果证明了本设计的有效性。

关键词：微带天线；全球定位系统；双频段；圆极化；A COMPACT DUAL-BAND GPS MICROSTRIP ANTENNAPENG xiang-fei, ZHONG Shun-shi, XU Sai-qing , WU Qiang（1.School of Communication and Information Engineering，Shanghai University,Shanghai 200072；2.Zhejiang Zhengyuan electric limited Company , Jiaxing Zhejiang 314003）Abstract: Recently in order to satisfy the demanded precision and reliability for the globe positioning system(GPS) , the dual-band circularly polarized(CP) is required. This paper describes thedesign of a probe-fed stacking two corner-truncated square microstrip patch antennas, which are using two different relative permittivity microwave ceramic substrates. Comparing with the conventional dual-band CP antenna with a same low relative permittivity and an air-gap layer between two patches, the size of this antenna is reduced and its fabrication is easier. Details of the proposed antenna design and experimental results are presented and discussed .The measured results confirm the validity of this design.Key words:microstrip antenna；GPS；dual-band ; circular polarization1 引言近年来微带天线由于它的尺寸小，成本低，易实现圆极化等优点在全球定位系统（GPS）应用中独占鳌头。

一种新型的圆极化贴片天线的研究

2007年全国微波毫米波会议论文集308 一种新型的圆极化贴片天线的研究张继龙卢春兰钱祖平（解放军理工大学通信工程学院，江苏南京，210007）摘要：本文研究了圆极化微带贴片天线，通过在普通圆形贴片开槽，提出了一种结构新颖的圆极化贴片天线。

仿真以及实测结果表明，该天线具有较宽的3dB波瓣和良好的圆极化性能，并且新型贴片天线的尺寸要小于普通的圆形或圆环形贴片天线的尺寸。

关键词：贴片天线；圆极化；轴比A Novel Circular-polarized MicrostripPatch AntennaZhang Ji-long Lu Chun-lan Qian Zu-Ping（Communication Engineering Institute of Science Technology University PLA, jiangsu nanjing,21007）Abstract: In this paper a novel circular-polarized microstrip patch antenna is given based on the study of common circular microstrip patch antenna. This new type of patch looks like common circular patch with some slots. Numerical results and measured data indicate that the new patch antenna has a wide beam and good performance of axial ratio. The radiation pattern of the antenna is very good. Another property of the new patch antenna is that the size of new patch antenna is smaller than common circular patch or annular patch antenna.Key word: patch antenna; circular-polarization; axial ratio1 引言*微带贴片天线由于重量轻、体积小、剖面低，此外还具有良好的方向性、灵活的馈电方式且容易与其他印刷电路集成等优点，在许多领域有着广泛的应用前景。

高次模对微带天线方向图的影响分析

高次模对微带天线方向图的影响分析作者：郑灵曹军陈嗣乔张小刚来源：《电子科学技术》2016年第06期摘要：微带天线具有剖面低、重量轻、易于与平台共形等优点，传统微带天线带宽较窄，目前已有一些方法来展宽微带天线的带宽，然而，设计中发现，展宽微带天线带宽的同时可能会导致高次模的出现，高次模的出现导致天线方向图的恶化，目前这方面研究还较少。

本文基于微带天线模型，分析了高次模对微带天线方向图的影响。

关键词：微带天线；高次模；方向图中图分类号：TN822 文献标识码：A 文章编号： 2095-8595（2016） 06-702-04电子科学技术 URL： http// DOI： 10.16453/j.issn.2095-8595.2016.06.008引言与其他天线相比，微带天线具有剖面低、重量轻、易于与平台表面共形、适于大批量生产等优点[1-5]，然而，传统微带天线带宽较窄，为了适应雷达、通信、电子对抗等系统的需求，目前国内外已提出了一些方法来展宽微带天线的带宽，如采用渐变结构、电阻加载、电容加载等 [6-11]。

然而，设计中发现，展宽微带天线带宽的同时可能会导致高次模的出现，高次模的出现导致天线方向图的恶化。

因此，研究高次模对天线方向图的影响具有重要意义，然而，目前这方面研究还较少。

本文基于微带天线模型，分析了高次模对微带天线方向图的影响。

1 物理模型微带天线物理模型如图1所示，天线宽度为a，长度为b，印刷在厚度为h1，相对介电常数为εr的微带板上，天线和反射板之间的距离h=h1+h2。

基于空腔模型，分析了a/b 值不同时天线的方向图，如图3所示，可以看到，随着天线宽度增加，天线方向图出现分裂。

a/b=0.05时，E面和H面最大辐射方向均在法向；a/b=0.5时，E面最大辐射方向仍在法向，H面方向图出现分裂；a/b=1时，E面和H面方向图均出现分裂。

由此可见，展宽微带天线带宽时，要防止因高次模的出现而导致天线方向图出现分裂。

一种具有宽角轴比特性的圆极化天线

一种具有宽角轴比特性的圆极化天线陶伟;赵玉军;王昕晔【摘要】分析了微带圆极化天线的工作原理,提出了一种双层电磁耦合型馈电结构以取代传统的馈电结构.通过优化L型微带馈电形状、馈线与辐射单元间的间距以及馈线阻抗匹配等措施,实现了一种具有宽角轴比特性的微带圆极化天线,并对其样机的增益、轴比和电压驻波比进行了测试,测试结果表明天线性能达到了预期研究目标.【期刊名称】《电波科学学报》【年(卷),期】2015(030)003【总页数】5页(P560-564)【关键词】轴比;圆极化;微带天线【作者】陶伟;赵玉军;王昕晔【作者单位】海军装备研究院,北京100161;陕西海通天线有限责任公司,陕西西安710075;海军工程大学电子工程学院,湖北武汉430033【正文语种】中文【中图分类】TN958.93引言随着卫星通信和全球卫星导航定位系统技术的发展,与之配套的圆极化天线凭借其独特的旋向正交特性,也得到了迅速发展与应用,有效提高了卫星通信和导航系统的接收灵敏度与定位精度,降低了系统接收误码率.实际工程应用中,为满足天线在上半空间具有均匀辐射特性和全向性等使用需求,要求圆极化天线具有较宽的覆盖波束,在更宽的波束范围有好的轴比特性.文献[1-3]表明,通过设计四臂螺旋、十字交叉振子和微带等天线形式均可实现宽波束的圆极化天线;其中微带圆极化天线因为具有体积小、重量轻、低剖面、低成本、易于实现量产和微带线路集成等众多优点,得到了更加广泛的关注和应用.但受天线底部金属腔和边界条件的影响,微带天线往往具有低仰角轴比特性差的缺点[4-5],严重阻碍了其实用化进程,因此如何提高微带圆极化天线的宽角轴比特性是当前天线领域的一个研究热点,也是难点.针对微带圆极化天线低仰角轴比特性差的问题,研究并设计了一种单馈点宽波束圆极化微带天线,采用L型微带馈电形状和耦合馈电的方式激励出正交方向的电场;通过调整馈电长度,实现辐射正交电场的90°相位差,进而实现天线的圆极化;重点在馈电结构上有所创新,采用双层电磁耦合型馈电结构取代传统馈电结构,使其同时具有一定的辐射功能,并在此基础上通过附加馈线阻抗匹配、调整馈线和辐射单元间的厚度等技术措施实现了指标的优化,完成了具有宽角轴比特性的微带天线设计.研制了天线原理样机并进行了测试,测试结果达到使用要求.1 微带圆极化天线基本理论与关键技术解决1.1 圆极化天线基本理论采用传输线模型理论分析微带天线,通过在导体贴片与接地板之间激励起电磁场,微带贴片可看作宽为W、长为L(一般L=λg/2)的一段微带传输线(λg为介质波长),其终端在L处呈现开路.对于圆极化天线,需要设计使其辐射出两个在空间互相垂直、幅度相等,相位相差90°的电磁波.通过微带天线的半波辐射结构,其在某一频率的谐振长度可由下式近似得出[6](1)式中: c是真空中光速; λg是介质波长； L是贴片的长度; εe是基板材料的等效介电常数; ΔL是等效贴片长度伸长,其值由以下公式计算:(2)(3)式中: H为基片厚度; εr是基片材料的相对介电常数; W为贴片宽度,在设计圆极化时与L近似相等.轴比是圆极化天线区别于其它天线的一个主要性能参数.任意极化波的瞬时电场矢量的端点轨迹为一椭圆,椭圆的长轴和短轴之比称之为轴比.轴比代表圆极化的纯度,是衡量整机对不同方向的信号增益差异性的重要指标,轴比小于3 dB的带宽定义为天线的圆极化带宽.在卫星通信和导航等应用中,由于卫星分布在上半球空间,而轴比易受到天线边界条件和外形结构等影响,所以保证天线在各个方向均有相近的敏感度非常重要.1.2 L型微带馈线与耦合馈电贴片结构设计技术所设计的宽角轴比圆极化天线采用了双层结构的电磁耦合型馈电,如图1所示.图中,h为介质基片厚度,L为近似正方形微带贴片的边长,S为L型微带馈线长边的长度,F为L型微带馈线的宽度.选择使用相同参数的介质基片1和介质基片2层叠,利用介质基片2将同轴探针馈电过渡到L型微带馈线,用L型微带馈线给位于介质基片1上的近似方形微带贴片间接馈电,通过介质基片1的地与介质基片2的贴片形成容性缝隙辐射电磁能量.L型微带馈线激励了两个空间上相互垂直、幅度近似相等、相位相差90°的电磁波.设计中在近似方形微带贴片上形成一个微调单元A,其作用近似一个并联的电容,对天线圆极化性能起到了优化作用.图1 采用双层电磁耦合馈电的微带右旋圆极化天线结构利用Ansoft HFSS电磁分析软件对天线进行了仿真,图2 给出了天线表面电流分布.从图中可以看出,电流主要分布在辐射贴片表面,但是在L型微带馈线表面还形成了少量电流分布,这部分电流将会产生辐射,由此表明,应用本设计可以通过调整微带馈线来调整其表面电流辐射,进而实现对整个天线方向图的调整,并最终达到调整方向图宽度以及天线宽角轴比的目的.图2 微带圆极化天线表面电流分布示意图为了验证微带馈线在上述设计中的调整作用进行了仿真计算.图3～5给出了不同馈线宽度对应的方向图、轴比(1.268 GHz的其中一个剖面)以及驻波比的仿真计算图示.从图中可以看出,馈线的宽度会影响天线波瓣宽度、阻抗匹配和电压驻波比,这充分证明了本设计中馈线对天线辐射的调节作用.图3 不同馈线宽度对天线方向图的影响图4 不同馈线宽度对天线轴比的影响图5 不同馈线宽度对天线电压驻波比的影响1.3 天线性能优化为获取最优的馈线宽度以实现好的低仰角轴比特性,主要从以下三方面进行优化. 1) 对天线的辐射性能进行优化.根据工作频率和工程需要的体积大小,选择合适的介质基片,采用厚度h=3.0 mm、介电常数εr=6.15的介质基片1和介质基片2进行小型化设计,增加介质厚度,增加辐射单元的电导,从而提高辐射效率;优化设计L型微带馈线的馈电方向和尺寸,实现右旋的圆极化和较宽的宽角轴比带宽,同时调整近似方形微带贴片和微调单元的尺寸,进一步提高圆极化性能.2) 进行带宽展宽设计.圆极化微带天线的工作带宽主要受限于阻抗带宽,所以附加馈线阻抗匹配可以有效展宽微带天线的工作带宽,通过增加馈线和辐射单元之间的厚度可增加阻抗带宽.通过在同轴馈电和L型微带馈线之间增加微带阻抗匹配段,增大天线的阻抗带宽和阻抗匹配调整自由度,使天线阻抗特性得到优化.3) 进行电压驻波比优化.增加介质基板厚度,将两层厚度为3 mm基片叠加为6 mm,增大了微带天线辐射的缝隙宽度,使得谐振腔中辐射出的能量增大.同时利用厚介质基片激励的表面波模式展宽了天线波束,提高了低仰角性能.通过以上设计,得到L波段内以1 268 MHz为中心频率的宽角轴比圆极化天线具体设计参数为:εr=6.15、h=3.0 mm、L=0.182λ0 mm、S=0.149λ0 mm以及馈线宽度F=0.076 λ0.2 宽角轴比圆极化天线的仿真测试图6～8是采用Ansoft HFSS电磁分析软件仿真的天线电气特性图.图6的仿真结果显示,在1 268±10 MHz频点的右旋圆极化最大增益分别为5.3 dBic、5.51 dBic、4.99 dBic,150°范围右旋圆极化最小增益分别为-0.1 dBic、-0.15 dBic、-0.33 dBic.图7的仿真结果显示,波束宽度150°范围轴比分别为2.01 dB、1.1 dB、3.1 dB. 图8仿真结果显示,1 268±10 MHz频段内天线的电压驻波比小于等于1.2;1268±30 MHz频段内天线的电压驻波比小于等于1.5.图6 右旋圆极化垂直面增益方向图图7 右旋圆极化轴比方向图图8 右旋圆极化电压驻波比3 样机测试结果在上述理论指导下,研制了这种宽角轴比圆极化天线原理样机(见图9).图9 宽角轴比微带圆极化天线实物照片根据工程使用要求在天线底部加装金属腔体后,使用128多探头球面近场天线测试系统进行了测试,结果如图10～12所示.3.1 天线增益测试结果由图10可以看出,中心频率1 268±10 MHz 内150°波束范围内圆极化增益大于等于-0.5 dBic实现了较宽波束范围内较高的辐射效率,满足卫星导航定位天线低仰角具有较高增益的要求.图10 实测垂直面增益方向图图11 实测轴比方向图3.2 天线轴比带宽测试结果由图11可以看出,中心频率1 268±10 MHz时,150°波束范围轴比小于等于3 dB,达到了预期的宽角轴比要求.3.3 天线电压驻波比测试结果由图12可以看出,在中心频率1 268±10 MHz范围内实测的天线电压驻波比小于等于1.2,天线在工作频率范围内具有较好的阻抗匹配特性.图12 实测电压驻波比4 结论设计具有宽角轴比特性的微带圆极化天线是当前卫星通信、卫星导航系统发展和应用的要求,具有很大挑战性.采用层叠结构的耦合贴片和L型微带馈线技术优化设计了右旋圆极化天线,研制了具有宽角轴比特性的圆极化天线样机,并对其增益、轴比和电压驻波比进行了测试.测试结果表明该天线在较宽的波束范围内具有好的轴比和极化增益,在更宽的频带内获得较低的电压驻波比,满足工程应用要求.参考文献[1] 白旭东, 曹岸杰, 唐晶晶, 等. 一种新型双频宽波束四臂螺旋天线的设计[J]. 中国电子科学院学报, 2002, l7(1): 81-84.BAI Xudong, CAO Anjie, TANG Jingjing, et al. Design of a novel dual-band wide-beam quadrifilar helix antenna[J]. Journal of China Academy ofElectronics and Information Technology, 2002, 17(1):81-84. (in Chinese) [2] 周进, 王玉峰, 常雷, 等. 一种宽波束圆极化十字阵子天线[J]. 通信对抗, 2014, 33(1): 40-45.ZHOU Jin, WANG Yufeng, CHANG Lei, et al. A wide beam circular polarization cross dipole antenna[J]. Communication Countermeasures, 2014, 33(1): 40-45. (in Chinese)[3] 陈玉林. 一种新型宽波束双圆极化天线的设计与仿真[J]. 火控雷达技术, 2014, 43(1): 94-97.CHEN Yulin. Design and simulation of a novel wide beam dual circular polarization antenna[J]. Fire Control Radar Technology, 2014, 43(1): 94-97. (in Chinese)[4] 罗远祉, 阳建平, 钟顺时. 圆极化微带天线的理论与实验[J]. 通信学报, 1989, 10(4): 20-27.LOU Yuanzhi, YANG Jianping, ZHONG Shunshi. Theory and experiment on circularly polarized microstrip antenna[J]. Journal of China Institute of Communications,1989, 10(4): 20-27. (in Chinese)[5] CHEN W S, WU C K, WONG K L. Novel compact circularly polarized square microstrip antenna[J]. IEEE Trans Antennas Propagat, 2001, 49(2): 340-342.[6] 林昌禄. 天线工程手册[M]. 北京: 电子工业出版社, 2002: 127.。

新型超宽带圆极化印刷天线

新型超宽带圆极化印刷天线李振亚;竺小松;张建华;刘汉【摘要】针对传统圆极化微带天线轴比带宽较窄的问题,提出了一种新型的超宽带圆极化印刷天线.该天线结构简单,仅由C型辐射贴片和改进后的地板组成.采用微带馈电模式,整个天线尺寸仅为25 mm×25 mm×1 mm.通过优化C型贴片和在地板上增加三角形和小长方形微扰结构,可以有效增加天线的阻抗带宽和轴比带宽.给出了天线的设计流程,从表面电流分布分析了圆极化天线的工作机理.加工了天线实物,并对其进行了测量.仿真和实测结果表明天线具有超宽的阻抗带宽和轴比带宽.天线的工作频带为4.4～11.6 GHz(相对带宽为90％),3 dB轴比带宽为3.6～11.3 GHz(相对带宽为103.4％).测量了天线的辐射性能和增益特性,实测结果与仿真结果吻合较好,证明了该天线的有效性.该天线可以应用于超宽带无线通信系统和卫星通信系统中.【期刊名称】《系统工程与电子技术》【年(卷),期】2019(041)001【总页数】5页(P9-13)【关键词】轴比;超宽带;圆极化;印刷天线【作者】李振亚;竺小松;张建华;刘汉【作者单位】国防科技大学电子对抗学院,安徽合肥230037;国防科技大学电子对抗学院,安徽合肥230037;国防科技大学电子对抗学院,安徽合肥230037;国防科技大学信息通信学院,湖北武汉430010【正文语种】中文【中图分类】TN820 引言近年来,随着现代无线通信和卫星导航通信的迅猛发展,圆极化天线的作用越来越凸显[1]。

圆极化天线相比线极化天线有其天然的优势,具有良好的多径反射能力和极化易匹配特性,抗干扰能力强,即使在恶劣的雨雾天气下也能较好的工作[2]。

传统的微带天线通常采用单点馈电和多点馈电法来实现圆极化性能,虽然能够实现圆极化特性,但是其轴比一般比较窄,加之微带天线本身的带宽也比较窄,很难应用于宽带圆极化天线系统中。

为了获得更大的系统容量,消除极化失配的影响,实现高数据传输,超宽带圆极化天线的设计成为了天线领域的热点研究方向[3-4]。

一种微带线馈电的宽带圆极化微带天线的设计

一种微带线馈电的宽带圆极化微带天线的设计尚玉玺;刘运林;何之煜【摘要】The broadband circularly polarized microstrip antenna with microstrip feeding is presented,which consists of mi⁃crostripfeeder,radiation patch and FR4 dielectric⁃slab. The requirements of microstrip antenna circularly polarized are realized by adding two annular slots which are located at the two opposite angles of rectangular slot on radiation patch. The axial⁃ratio bandwidth of the antenna is improved effectively by adjusting the size of microstrip feeder. The axial⁃ratio bandwidth can reach 43.8%(2.5~3.9 GHz).%提出一种微带线馈电的宽带圆极化微带天线，它由微带馈线、辐射贴片和FR4介质板组成，在辐射贴片的矩形槽对角添加两个环形结构，实现了微带天线圆极化的要求，通过调整微带馈线的尺寸，有效改善了天线的轴比带宽。

该天线单元的轴比带宽达到了43.8%（2.5~3.9 GHz）。

【期刊名称】《现代电子技术》【年(卷),期】2015(000)013【总页数】4页(P67-70)【关键词】宽带;圆极化;微带线馈电;微带天线【作者】尚玉玺;刘运林;何之煜【作者单位】西南交通大学电磁场与微波研究所，四川成都 610031;西南交通大学电磁场与微波研究所，四川成都 610031;西南交通大学电磁场与微波研究所，四川成都 610031【正文语种】中文【中图分类】TN92-34现代无线通信系统对天线的性能要求越来越高，单纯线极化天线已无法满足要求，因此圆极化天线的应用越来越广泛。

蛇形单极子天线匹配系统设计优化研究

蛇形单极子天线匹配系统设计优化研究杨晓昆; 张正平【期刊名称】《《通信技术》》【年(卷),期】2019(052)003【总页数】6页(P724-729)【关键词】单极子天线; 输入匹配; 集总RLC; 传输模块【作者】杨晓昆; 张正平【作者单位】贵州大学大数据与信息工程学院贵州贵阳 550025【正文语种】中文【中图分类】TN920 引言自偶极子天线诞生以来，它凭借优秀的方向增益性能，在众多商业化产品中得到了广泛应用。

但是，随着科学技术的发展和生活水平的提高，人们也从最开始的满足于产品的功能向追求产品性能的方向转变[1］。

此时，传统自身特性阻抗73.2 Ω、方向增益2.15 dB的半波偶极子天线，因其固有的1/2波长特性尺寸，已经无法满足消费者的诸多需求。

为了提高无线产品适应性，单极子天线以其高方向（2.15 dB+3 dB）[2］性和只需1/4波长的便携性等优点被提出，迅速得到了业内的广泛认可，并开始应用于越来越多的商业产品中，如最初的无线电收音机等。

随着无线电技术的迅猛发展，需要接入的无线通信可携带式设备越来越多样化，也令ISM频段的可用区间愈发匮乏。

为了在高集成的PCB中在保证功能不变的情况下能够继续优化天线尺寸，倒L型天线（ILA）被提出。

这款天线因其贴片和可折叠优势，不仅保留了单极子天线的各种基本特性，而且以ILA天线为蓝本的各类多适应性天线层出不穷[3］。

如图1所示，几种贴片天线均以倒L型天线的设计思路为基础，并结合自身功能需求，衍生出了几种天线类型。

图1 ILA衍生天线如图1所示，微型收音机中的ILA天线，根据产品需求、结构重构等进行了一系列设计。

其中，如图1（1）所示天线，可以同时有效地接收环境中2.4 GHz/5.8 GHz[4］的双频段电磁能量；如图1（2）中所示的天线设计，目的在于能够令天线在50 Ω处实现天线完全匹配；如图1（3）所示的蛇形天线，因其结构特点，可以在868 MHz频段内对电磁信号有着良好的吸收特性，适用于多种模块化传输固件。

一款双频双宽带双圆极化Fabry

现代电子技术Modern Electronics Technique2023年11月1日第46卷第21期Nov. 2023Vol. 46 No. 210 引言随着信息技术的迅猛发展，空天地海一体化通信系统已经成为未来通信网络的发展趋势，卫星通信具有覆盖范围广、不受地理条件限制等优点，其作为天基通信的重要组成部分和未来6G 网络技术发展的重要方向，已经成为学术界研究的热点[1]。

天线作为卫星通信系统的关键组成部件之一，要求具有宽带宽、高增益、圆极化和结构简单、易于集成等特性。

由于法布里⁃珀罗（Fabry Perot, FP ）谐振腔天线具有增益高、馈电简单的特性，自从其诞生以来便受到了学术界的广泛关注[2]。

FP 天线具有馈电结构简单、增益高的优点，近年来学术界对于FP 天线的双频段工作、宽带性能以及如何实现圆极化辐射做了大量研究工作，其在卫星通信系统中有良好的应用潜力。

对于FP 天线的双频工作特性[3⁃5]，可采用频率选择表面（Frequency Selective一款双频双宽带双圆极化Fabry⁃Perot 谐振腔天线吕军1，钟选明2（1.国能包神铁路有限责任公司，内蒙古鄂尔多斯 017000；2.成都交大运达电气有限公司，四川成都 610000）摘要：文中设计了一款双频双宽带双圆极化的法布里⁃珀罗（FP ）谐振腔天线。

传统的FP 天线具有高增益特性但是难以实现宽带及双频带工作，为了改善其性能，提出一种具有双频正相位梯度的部分反射表面，利用其正相位梯度特性弥补电磁波频率升高带来的空间相位变化，从而在较宽的带宽内满足FP 天线的谐振条件以实现宽带辐射。

通过加载寄生贴片以及缝隙耦合馈电的方式设计宽带圆极化馈源，并且采用人工磁导体结构替代传统的金属地板，在同一谐振腔高度下满足两个频段的谐振条件，简化了双频FP 天线的结构。

全波仿真结果表明，所提出的FP 天线3 dB 轴比带宽分别为10.1%和13.8%，峰值增益达到12.45 dBi 和11.9 dBi ，3 dB 增益带宽分别为11.5%和14.8%。

新型L频段双圆极化微带阵列天线的设计

新型L频段双圆极化微带阵列天线的设计李文;姚宜东;徐毅;袁伟涛;杨新华;王启申【摘要】Circularly-polarized array antennas attract more and more attentions in the modern wireless applications because of its specific performance characteristics. A L-band circularly-polarized microstrip patch antenna working in wide axial ratio bandwidth is proposed. The antenna adopts the special double feed network, thus to provide 0 degree feed and 90 degree feed to the two adjacent sides of radiation patch respectively. Two layers of feed network are same in structural size, and connected through the bridge to ensure that the two adjacent sides of radiation patch have 90 degrees phase difference, thus improving circular polarization performance of the antenna. The simulation results show that the microstrip array antenna could work at 1.525~1.559 GHz; with double circular polarized antenna; antenna gain> 13 dBi; VSWR<1.5; E and H plane lobe width> 25°.%圆极化阵列天线由于其自身的性能特点，在现代无线应用中越来越受到广泛的关注。

双频带圆极化微带阵列天线设计

双频带圆极化微带阵列天线设计胡金艳;杨君;秦文华;赵建平;徐娟【摘要】利用旋转馈电技术设计了一种双频带圆极化微带阵列天线,以扩充天线的通信容量,提高抗干扰能力.天线由四个对角切角的矩形贴片和一个金属矩形环组成.天线利用贴片切角实现圆极化,利用两个贴片的对角线长度不等实现双频特性.天线中心的矩形环既可当做馈电网络,为圆极化波提供所需的递增相位,又可以提高天线的辐射性能.最后,利用电磁仿真软件HFSS对天线的性能进行数值计算,阵列天线的-10 dB阻抗带宽分别为1.3～1.4 GHz和1.55～1.58 GHz,3 dB轴比带宽分别为1.36～1.42 GHz和1.6～1.62 GHz.%A dual-band circularly-polarized micro-strip patch array, by using a sequential-phase feeding network, is designed and implemented, thus to expand the communication capacity and improve the anti-interference capability. The antenna, composed of four rectangular patches with diagonally tangential angles and one metal rectangular ring, is circularly-polarized by using the corner patch, while the dual-frequency is realized by using the unequal diagonal lengths of two patches. The mental square ring in the center of the antenna may act as a feeding network, which provides both the increasing phase for circularly-polarized wave and a radiator to enhance the performance of the antenna. Finally, the simulation on the antenna with HFSS software indicates that the -10 dB impedance bandwidth of the patch array is 1.3～1.4 GHz and 1.55～1.58 GHz , and the measured 3dB AR bandwidth 1.36～1.42 GHz and 1.6～1.62 GHz respectively.【期刊名称】《通信技术》【年(卷),期】2018(051)001【总页数】6页(P234-239)【关键词】旋转馈电;双频带;圆极化;微带阵列【作者】胡金艳;杨君;秦文华;赵建平;徐娟【作者单位】曲阜师范大学物理工程学院,山东曲阜 273165;曲阜师范大学物理工程学院,山东曲阜 273165;曲阜师范大学物理工程学院,山东曲阜 273165;曲阜师范大学物理工程学院,山东曲阜 273165;曲阜师范大学物理工程学院,山东曲阜273165【正文语种】中文0 引言与线极化天线相比，圆极化天线有几个重要的优势：对抗多径干扰或衰落﹑减少电离层的“法拉第旋转”效应和降低极化失配。

一种双频嵌套卫星通信天线设计

2
天线设计
用微带线的形式，分别布置了三个威尔金森功分
2.1
天线结构设计
器［10］，实现了四端口功率平分，中间设计有隔离电
本文所设计的双频嵌套卫星通信天线结构如
图 1。天线的尺寸为Φ140mm×430mm，天线分为辐
阻 100Ω，相位差依次 90°。两副馈电网络仿真模型
图如图 3 所示。
射部分、馈电网络和支撑骨架三部分，其中天线辐
requirements.
double-frequency，nested，power distribution phase shifting feed network
Key Words
TN828.5
Class Number
1
性要求［8］，因此被广泛地运用于卫星通信。螺旋天
引言
线由金属导体按螺旋状绕制而成的，根据螺旋半径
的，同时在轴向上能够缩短天线的尺寸，为了保证
自的工作频段要求分别使用开路与短路的不同工
作方式。内部天线工作频率高、工作带宽较宽对
天线的 0dB 波束宽度，天线采用四分之三波长的电
尺寸。收信天线频采用 3/4 波长开路天线
不影响内部天线的性能，该天线采用二分之一波长
作者简介：毛勇，男，硕士，高级工程师，研究方向：舰船通信技术。苏赫，男，硕士，工程师，研究方向：电磁场与微波技
术。杨懿楠，女，硕士，工程师，研究方向：电磁场与微波技术。罗曼，女，硕士，研究员，研究方向：舰船通信技术。
2021 年第 1 期
35
舰船电子工程
为 0°、90°、180°、270°。
tenna is used to reduce the antenna volume, expand the antenna frequency band, and the receiving and transmitting antennas are

北斗高精度测量型天线的研究

北斗高精度测量型天线的研究吴多龙;周梓发;李瑞;鲍志雄【摘要】针对北斗高精度测量的需求，要求接收天线在北斗卫星定位导航系统的两个频段上均能良好工作。

介绍了一种有源双层微带天线，馈电网络由六个电桥结合八个探针共同组成，实现了在宽角度范围内的右旋圆极化特性，保证了相位中心的稳定度。

接收到的北斗卫星信号通过两级低噪声放大器进行放大后滤波，既保证了接收机信号的增益要求，又提高了带外噪声的抑制能力。

经仿真和实验测试结果对比表明：该天线能够充分满足北斗高精度测量应用的要求。

%On the basis of requirements with high-precision measurements for Com- pass Navigation Satellite System（CNSS）, antennas work well at B1 （1561 MHz） and B2 （1207 MHz）bands. In this paper,a novel dual-layer microstrip active antenna is proposed and its feeding net is composed of six couplers and eight-feed points, which can better realize right-hand circular polarization （RHCP）characteristics in wide angles and stabilize the phase center. The signal received from compass satellites is amplified by dual low-noise amplifiers and then filtered, which ensures requirements of receiver gain in bands and improves out-of-band noise suppression. Simulated and measured results show that the antenna meets the requirements of high-precision measurement applications.【期刊名称】《电波科学学报》【年(卷),期】2011(026)005【总页数】5页(P1008-1012)【关键词】北斗卫星定位导航系统;双层微带天线;馈电网络;高精度测量;相位中心;右旋圆极化【作者】吴多龙;周梓发;李瑞;鲍志雄【作者单位】广东工业大学物理与光电工程学院,广东广州510006;广东工业大学物理与光电工程学院,广东广州510006;广东工业大学物理与光电工程学院,广东广州510006;广州市中海达测绘仪器有限公司,广东广州511400【正文语种】中文【中图分类】TN8211.引言利用卫星定位系统进行高精度测量在测绘领域具有广泛的应用，但是对天线的性能提出了更高的要求。

频率可重构的单极子天线设计

频率可重构的单极子天线设计成根;段美玲;李文妮;韩丽萍【摘要】本文设计了一种频率可重构的单极子天线.天线由一个阶梯型馈线、两个L型枝节和一个矩形接地板组成.两个理想开关加载在馈线与枝节之间,通过控制开关状态改变天线的表面电流分布,从而实现频率可重构.天线的尺寸为35 mm×40 mm.仿真和测量结果表明:该天线可以在两个单频模式(2.4 GHz和5.2 GHz)以及一个双频模式(2.4 GHz/5.2 GHz)之间切换.天线在不同模式下都具有稳定的辐射方向图.%This paper presents a frequency-reconfigurable monoole antenna.The antenna consist of a stepped microstrip feed line,two L-shaped stubs and a rectangular ground plane,and two ideal switches are loaded between microstrip feed line and stubs.By changing the states of the switches,the current distribution will be altered,and then frequency reconfigurability is achieved.The overall size of the antenna is 35 mm × 40 mm.The simulated and measured results show that the proposed antenna can operate at two single-band modes (2.4 GHzand 5.2 GHz) and a dual-band modes (2.4 GHz/5.2 GHz).The proposed antenna has a stable radiation patterns in different modes.【期刊名称】《测试技术学报》【年(卷),期】2017(031)002【总页数】5页(P148-152)【关键词】阶梯型馈线;频率可重构;单频模式;双频模式;单极子天线【作者】成根;段美玲;李文妮;韩丽萍【作者单位】山西大学物理电子工程学院,山西太原030006;山西大学外国语学院,山西太原030006;山西大学物理电子工程学院,山西太原030006;山西大学物理电子工程学院,山西太原030006【正文语种】中文【中图分类】TN821+.3近年来，随着无线通信技术的快速发展，对移动设备工作在不同应用环境的需求日益增加. 可重构天线依据设备所处应用环境，可以对天线的工作频率、辐射方向图、极化等特性进行重构，并且具有成本低、功能多、体积小和易于集成等优点，引起了国内外学者的广泛关注.频率可重构天线通过加载开关元件，如PIN二极管[1]，变容二极管[2]，射频微机电系统(RF-MEMS)[3]，以及GaAs场效应晶体管(FET)[4]等，改变天线的表面电流分布，从而改变天线的谐振频率，实现频率可重构. 然而，许多频率可重构天线只能工作在单频带或双频带模式. 文献[5]在半圆形贴片上的调谐枝节中加载3个PIN二极管，天线可以在4个单频带之间切换，频比为1.7∶1. 文献[6]在L型缝隙中加载5个PIN二极管，天线工作在6个单频带模式，工作频带为2.2～4.75 GHz. 此外，有学者在S型[7]、 T型[8]缝隙中加载PIN二极管，通过控制二极管的状态改变缝隙长度，实现频率可重构. 文献[9]在两个贴片单元分别刻蚀C型缝隙，并在馈电网络与两个贴片单元的连接处加载一对PIN二极管，天线工作在两个双频带模式和一个宽频带模式. 文献[10]设计了一个双频带可重构的缝隙天线，在缝隙的适当位置加载两个变容二极管，通过控制变容二极管的偏置电压，天线可以实现双频带的切换.本文设计了一种频率可重构的单极子天线. 在馈线与两个L型枝节的连接处分别加载一个开关，通过控制开关的通断实现频率可重构. 仿真和测量结果表明天线可以工作在两个单频模式和一个双频模式.天线的结构如图 1 所示. 该天线包括3层，上层为馈线和辐射单元，中间层为介质基板，下层为接地板. 辐射单元中两个L型枝节分别实现不同的工作频率，阶梯型馈线改善天线的阻抗匹配. 通过在馈线与两个枝节的连接处分别加载一个开关并控制二极管的通断，改变天线的表面电流分布，实现频率可重构. 设计的天线工作频率为2.4 GHz和5.2 GHz，利用三维电磁仿真软件HFSS进行仿真. 选用相对介电常数为4.4，厚度为1.6 mm的FR4介质基板，优化的参数为： W=35 mm, L=40 mm, lf=11 mm, wf =3 mm, lp=7 mm, wp=6 mm, l1=3.7 mm,l2=4.8 mm, l3=16.2 mm, l4=8.5 mm, d=2 mm, lg=14 mm, wg=40 mm. 理想开关用尺寸为1 mm × 0.8 mm 的铜片代替，用铜片的有无表示开关的导通和断开. 表1给出了天线的工作模式.图 2 为天线的仿真反射系数曲线. 由图2可知，在模式1(S1导通， S2断开)和模式2(S1断开， S2导通)时，天线工作在单频模式，谐振频率分别为f1=2.4 GHz 和f2=5.2 GHz；在模式3(S1导通， S2导通)时，天线工作在双频模式，谐振频率为f1/f2=2.4/5.2 GHz.为了说明天线的工作原理，对天线3种工作模式的表面电流分布进行了研究，如图3 所示. 图3(a)和图3(c)给出了天线模式1和模式3谐振频率为f1的电流分布，由图可知，电流主要集中在较长的L型枝节. 在模式2和模式3，谐振频率f2的电流主要集中在较短的L型枝节和阶梯型馈线边缘，如图3(b) 和3(d)所示.通过对天线进行敏感性分析，发现阶梯型馈线主要影响天线的匹配性能， L型枝节长度影响天线的谐振频率. 在分析某一参数对天线性能的影响时，其它参数均保持不变. 图 4 和图 5 分别给出了在模式3情况下阶梯型馈线长度lp和宽度wp对天线反射系数的影响. 从图4中可以看出，随着lp的增加，天线高频段的阻抗匹配逐渐变好. 由图 5 可知，采用普通微带馈线(wp=wf=3 mm)时，天线高频部分阻抗匹配很差，采用阶梯型馈线时高频段的匹配性能得到明显改善. 图 6 给出在模式3情况下l1和l2变化时的反射系数曲线. 从图中可以看出，随着l1或l2的增加，谐振频率f2逐渐降低，而f1保持不变. 图 7 为在模式3的情况下， l3和l4对天线反射系数的影响. 由图可知，随着l3或l4的减少，谐振频率f1逐渐变大，而f2基本不变.天线印制在相对介电常数为4.4的FR4介质基板上，图 8 为天线的实物图. 采用Agilent公司N5230A矢量网络分析仪测量天线的反射系数，采用Lab-Volt公司8092型自动天线测量系统测量天线的方向图和增益.图 9 为天线仿真和测量的反射系数曲线. 天线的测量结果和仿真结果基本吻合. 从图中可以看出，天线能够实现3个不同的工作模式： 2.4 GHz和5.2 GHz两个单频模式以及2.4 GHz/5.2 GHz一个双频模式，每个模式的-10 dB带宽分别为9.5% (2.31～2.54 GHz)，9.1% (4.95～5.42 GHz)以及10.5% (2.23～2.48GHz)/12.6% (4.87～5.41 GHz). 仿真和测量结果的差异主要是由介质基板介电常数的偏差以及制作误差引起.图 10 是天线仿真和测量的归一化辐射方向图. 从图中可以看出，测量结果与仿真结果基本一致. H面的方向图基本是全向型， E面的方向图基本呈“8”字型. 天线在模式1，模式2和模式3的峰值增益分别为2.86, 1.91,以及1.67/2.45 dBi，满足无线通信系统的要求.本文设计了一种频率可重构的单极子天线. 通过在馈线与两个L型枝节的连接处分别加载一个开关，控制开关的状态，选择相应的辐射单元，实现频率可重构. 天线在各个模式具有稳定的辐射性能并且结构简单容易制作，可以满足Bluetooth，WLAN等无线通信系统使用.【相关文献】[1] Majid H A, Abdul Rahim M K, Hamid M R, et al. Frequency-reconfigurable microstrip patch-slot antenna[J]. IEEE Antennas and Wireless Propagation Letters, 2013, 12: 218-220.[2] Gu H, Wang J P, Ge L. Circularly polarized patch antenna with frequency reconfiguration[J]. IEEE Antennas and Wireless Propagation Letters, 2015, 14: 1170-1173.[3] Zohur A, Mopidevi H, Rodrigo D, et al. RF MEMS reconfigurable two-band antenna[J]. IEEE Antennas and Wireless Propagation Letters, 2013, 12: 72-75.[4] Yang X L, Lin J C, Chen G, et al. Frequency reconfigurable antenna for wireless communications using GaAs FET switch[J]. IEEE Antennas and Wireless Propagation Letters, 2015, 14: 807-810.[5] Desmond Sim C Y, Han T Y, Liao Y J. A frequency reconfigurable half annular ring slot antenna design[J]. IEEE Transactions on Antennas and Propagation, 2014, 62: 3428-3431.[6] Majid H A, Rahim M K A, Hamid M R, et al. A compact frequency-reconfigurable narrowband microstrip slot antenna[J]. IEEE Antennas and Wireless Propagation Letters, 2012, 11: 616-619.[7] Peroulis D, Sarabandi K, Katehi L P B. Design of reconfigurable slot antennas[J]. IEEE Transactions on Antennas and Propagation, 2005, 53: 645-654.[8] Anagnostou D E, Gheethan A A. A coplanar reconfigurable folded slot antenna without bias network for WLAN applications[J]. IEEE Antennas and Wireless Propagation Letters, 2009, 8: 1057-1060.[9] Abutarboush H F, Nilavalan R, Cheung S W, et al. A reconfigurable wideband and multiband antenna using dual-patch elements for compact wireless devices[J]. IEEE Transactions on Antennas and Propagation, 2012, 60: 36-43.[10] Behdad N, Sarabandi K. Dual-band reconfigurable antenna with a very wide tunability range[J]. 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A Novel Dual-Band Circularly-Polarized Wide-BeamQuadrifilar Helix AntennaXudong Bai, Di Yang, Jingjing Tang, Junping Geng, Ronghong Jin, Xianling Liang* Department of Electronic Engineering, Shanghai Jiao Tong University, Shanghai, 200240, P. R. ChinaE-mail:baixudong@Abstract—The satellite communication systems and satellite navigation systems usually require the antenna to provide compact size, wide beam, circular polarization and high gain at low elevation. Based on these needs, a new type of dual-band wide beamwidth quadrifilar helix antenna (QHA) is designed. The available dual-band is achieved by putting two similar antenna structures operated at different frequency bands in an inner and outer coaxial way, and fed by a split coax balun. A 90° self-phased structure is formed through the quadrifilar helix characteristic to obtain a wide beam and good circular polarization performance. Experimental results of the fabricated antenna show that, the measured results are consistent with the simulated ones, and the antenna can provide good wide-beam circular polarization characteristics in both frequency bands.I.INTRODUCTIONWith the development of GPS and SNS, antennas with dual-band even triple-band are required for many applications. But it is difficult for quadrifilar helix antenna (QHA) to cover the multi-band because of its structural features. Planar spiral antenna can get a particularly wide frequency band due to the use of self-complementary structure [1]. As for the single-arm helix antenna, the bandwidth can be enhanced by increasing or decreasing the pitch angle. Resonant QHA, because of its narrow band, cannot use the regulation method of adjusting the pitch angle to achieve the dual-band or multi-band.References [2-4] use several arms resonant at different frequencies instead of a single one to obtain the dual-band or multi-band, and get a 90° phase shift through the feed network; but this structure has the following limitations: Firstly, when the operation frequency is very low, the feed network needs a large space, thus the antenna will be seriously restricted by the reference ground. Secondly, when the several operation bands are farther apart, the feed network is difficult to achieve the 90° phase shift in a relatively wide band range. Meanwhile, since the length gap of the radiation arms resonant at different frequencies is very large; it is hard to find a suitable combination of the arms.In this paper, a novel dual-band wide-beam quadrifilar helix antenna for satellite communication and navigation systems is designed, the available dual-band is achieved by putting two similar antenna structures operated at different frequency bands in an inner and outer coaxial way, and the antenna is fed by a split coax balun. A 90° self-phased structure is formed through the quadrifilar helix characteristic to obtain a wide beam and good circular polarization performance.II.ANTENNA STRUCTUREFig.1 shows the configuration of the proposed antenna. Two similar antenna structures operated at different frequency bands are fixed in an inner and outer coaxial way. The antenna is feeded by the split coax balun, the λ/4 long balun slots isolate opposite elements from one another at the feed point. Both ends of the helical elements are shorted to the sheath of the balun, a 90° self-phased structure is formed through the quadrifilar helix characteristic to obtain a wide beam and good circular polarization performance. To obtain the 90° phase shift in the currents, the elements of one bifilar helix are adjusted longer than resonance, while the other is adjusted shorter. The resonance length of each volute is λ/2, and the antenna is operating in UHF band and S-band respectively. The outer structure is resonant in the UHF band, and each volute is 1/2 turn; the axial length of the outer volutes is L=68.8mm, and the radius of one outer bifilar helix is D1=36mm, the other is D2=36mm. The inner structure is resonant in the S-band, and each volute is 3/4 turn; the axial length of the inner volutes is l=42.8mm, and the radius of one inner bifilar helix is d1=23mm, the other is d2=22mm.(a) Side view (b) Top view(c) 3D-viewFig.1 Geometry of proposed antenna.978-1-4673-0462-7/12/$31.00 ©2012 IEEEIII. RESULTS AND DISCUSSIONTo verify the proposed antenna design, a prototype antenna is fabricated, as shown in Fig.2. The simulated and measured return loss versus frequency is shown in Fig.3. It is observed that the measured return loss is below -10dB in both frequency bands (<-11 dB in UHF-band and <-15 dB in S-band). Fig.4 gives the measured and simulated radiation patterns at the center frequencies of both bands, the antenna shows a wide angular coverage of more than 130°, and the gains at both bands are about 5 dBi. The measured and simulated AR is plotted in Fig.5. It shows that the AR is below 3dB when theta varies from -67° to 70° at the center frequency of UHF band, and the AR is below 3dB when theta vary from -62° to 58° at the center frequency of S-band.Fig.2 Photograph of the test QHAFig.3 Simulated and measured reflection coefficient.(a) Center frequency in UHF band(b) Center frequency in S-bandFig.4 Measured and simulated radiation patterns(a) Center frequency in UHF band(b) Center frequency in S-bandFig.5 Measured and simulated axial ratioIV. CONCLUSIONA novel dual-band wide-beam quadrifilar helix antenna for satellite communication and navigation systems is presented in this paper. The antenna has a good impedance bandwidth (S 11<-10dB) in both frequency bands. The antenna gains are about 5dBi in both frequency bands, and the 3dB beamwidth is more than 130°. On the whole, the antenna can provide good wide-beam circular polarization characteristics in both frequency bands.A CKNOWLEDGEMENTThis work was supported by “973” (2009CB320403), the Natural Science Foundation of Shanghai under Grant (10ZR1416600), the Doctoral Fund of Ministry of Education of China under Grant (20090073120033), the National Science and Tec hnology Major Project of the Ministry of Science (2011ZX03001-007-03), the National Science Fund for Creative Research Groups under Grant (60821062) and the Scientific Research F oundation for the Returned Overseas Chinese Scholars, State E ducation Ministry.R EFERENCES[1] Jia Lao, Ronghong Jin and Junping Geng, “An Equiangular SpiralAntenna with Parabolic Back-Cavity,” Microwave and Optical Technology Letters. 2008.[2] WenYi Qin, JingHui Qiu, Qi Wang, “A Novel Multi-frequencyQuadrifilar Helix Antenna,” IEEE Antennas and Propagation Society international Symposium. 2005(1B): 467-470.[3] M. Hosseini, i, M. Hakkak. “Design and Implementation of A Dual-bandQuadrifilar Helix Antenna,” IEEE 10th Int. Conf. on Mathematical Methods in Electromagnetic Theory. 2004, 9:493-495.[4] M. Hosseini, i, M. Hakkak, P. Rezaei. “Design of a Dual-BandQuadri ﬁlar Helix Antenna,” IEEE Antennas and Wireless Propagation Letters. 2005(4):39-42.。