input impedance matching with fully differential amplifiers

1By:Serial #:Operations Manual Portico 517500 Series DI / Pre / Comp2Portico 517: 500 Series DI / Pre / Compressor User GuideThank you for your purchase of the 517: 500 Series DI / Pre / Compressor. Everyone at Rupert Neve De-signs hope you enjoy using this tool as much as we have enjoyed designing and building it. Please take note of the following list of safety concerns and power requirements before the use of this or any Portico Series product.SafetyIt’s usual to provide a list of “do’s and don’ts” under this heading but mostly these amount to common sense issues. However here are some reminders:Don’t operate your Portico™ module in or around water! Electronic equipment and liquids are not good friends. If any liquid is spilled such as soda, coffee, alcoholic or other drink, the sugars and acids will have a very detrimental effect. Sugar crystals act like little rectifiers and can produce noise (crackles, etc.). SWITCH OFF IMMEDIATELY because once current starts to flow, the mixture hardens, can get very hot (burnt toffee!) and cause permanent and costly damage. Please contact support as soon as possible *************************************.Don’t be tempted to operate a Portico module with the cover removed. The cover provides magnetic screening from hum and R.F. stray fields.Power RequirementsEach Portico 517 is fitted for use with standard 500 Series Rack Mounts and requires 110-125 mAmps @+/- 16VMIC PRE0-66dBINTERNAL JUMPER SETTINGSOPTION A (FACTORY DEFAULT)FAST - 5ms/50msOPTION BSLOW - 250ms/500ms COMPRESSOR TIME CONSTANTOPTION A (FACTORY DEFAULT)OUT OPTION B IN - 80HzHIGH PASS FILTERGain +66dB in 6dB incrementsCompressor Single knob opto-coupler compressor with 2:1 ratioBlend Determines balanceof Mic and DI inputsInstrument Gain From 0 to 30dB of gain on instrument input 48V48V Phantom Power provided by 500 series rackSilkSilk adds nostalgic warmthand presence to when engaged Instrument InputFront panel TRS Input and passive through for hi-z sourcesPortico 517: Front Panel3MICROPHONE PREAMPLIFIER DESIGN NOTESIn former years, before the introduction of solid state amplifiers, transformers were necessary tostep up to the very high input impedance of tubes, and to provide a balanced input for the microphone line. An input impedance of 1,000 or 1,200 ohms became established for microphones having a source impedance of 150 or 200 ohms, with connection being made on a twisted twin screened cable (This type of cable, while excellent for low impedance work, has high capacitance between its conductors and between each conductor and screen. Resultant high frequency lossesare excessive with piezo pickups and may cause resonances with magnetic pickups.) Thus microphones were not heavily loaded. Condenser microphones worked off high voltage supplies(300V!) on the studio floor which polarized the diaphragms and powered a built-in pre-amplifier.More and more microphones were needed as “Pop” music gained ground and this led to the popular and efficient method of 48-volt “Phantom” powering that was built into the multi-channel recording Console – in place of numerous bulky supplies littering the studio, a miniature pre-amplifier nowbeing fitted inside the microphone casing.The 48-volt supply was fed to the microphone through balancing resistors so it was impossible forthis voltage to actually reach the microphone, resulting in low polarizing volts and virtual starvation of the little pre-amp inside the microphone. Nevertheless amazingly good microphones were designed and made, becoming the familiar product we use today. If a low value resistive load is connected tothe output of an amplifier, that amplifier has to produce power in order to maintain a voltage across that load. Obviously if we want more voltage (output from the microphone) we need to provide a larger supply for the amplifier or settle for a lighter load. A microphone is a voltage generator, not a power amplifier. Most microphones give their most accurate performance when they are not loadedby the input impedance of a traditional preamplifier. If the microphone uses an electronic circuit (transformerless) output, a low value of load impedance can possibly stress the little microphone pre-amplifier, causing slew rate and compression at high levels.On the other hand, a high value of load impedance allows the microphone to “breathe” and give ofits best, this being particularly advantageous with very high level percussive sounds. If the microphone has an inductive source (such as would be the case if it has a transformer output) a low value of load impedance causes the high frequencies to roll off due to leakage inductance in the transformer in addition to the above amplifier distortion (This can be an advantage with some microphones!).For this reason we have provided a high value of input impedance that will load microphones to the smallest possible extent and makes the best possible use of that limited “Phantom” 48-volts supply. DYNAMIC RANGETraditionally, high quality microphones such as ribbons, had very low source impedances – as low as 30 ohms at the output of a ribbon matching transformer. Moving coil microphones were higher but had not been standardized as they are today. Condenser microphones, before the days of semiconductors, used tube head amplifiers that were coupled to the outgoing line with a transformer. Microphone amplifiers, such as in a mixing console, also used tubes and these typically have a high input impedance. Microphones are Voltage generators, not Power generators. It is always desirable to deliver the maximum possible signal voltage into the amplifier. It was traditional to provide an amplifier input impedance of about 1,000 or 1,200 ohms; about 5 or 6 times the source impedance of the microphone. This provided relatively low loading on the microphone – whatever its type – and went a long way to avoid voltage loss.4In the early 1960’s when the “Pop” music scene was exploding and sound levels in the Studio became very high, there was concern that the head amplifiers in Condenser microphones would overload if the Console input impedance was too low. In the early days of Consoles I was asked to provide higher input impedance than the normal 1,000 ohms. This of course, resulted in less “step-up” in the Console input transformer and there were then fears that we would lose out at the other end of the scale; Noise. The fact that microphones were less heavily loaded allowed an increased microphone signal. The reduced loading also resulted in less deviation of frequency response due to variation of microphone impedance and consequently less distortion at high levels.The Portico 517 microphone amplifier provides an input impedance of 10,000 ohms which means that variations in microphone source impedance with frequency, have only a very small effect on the sonic quality. This high input impedance has minimal effect on microphone output and loading with the result that microphone distortion is very low adding up to a noticeable improvement in “transparency”.A NOTE ON DISTORTIONThe human hearing system is a remarkably complex mechanism and we seem to be learning more details about its workings all the time. For example, Oohashi demonstrated that arbitrarily filtering out ultrasonic information that is generally considered above our hearing range had a measurable effect on listener’s electroencephalo-grams. Kunchur describes several demonstrations that have shown that our hearing is capable of approximately twice the timing resolution than a limit of 20 kHz might imply(F=1/T or T=1/F). His peer reviewed papers demonstrated that we can hear timing resolution at approximately with 5 microsecond resolution (20 kHz implies a 9 microsecond temporal resolution, while a CD at 44.1k sample rate has a best-case temporal resolution of 23 microseconds).It is also well understood that we can perceive steady tones even when buried under 20 to 30 dB of noise. And we know that most gain stages exhibit rising distortion at higher frequencies, including more IM distortion. One common IM test is to mix 19 kHz and 20 kHz sine waves, send them through a device and then measure how much 1 kHz is generated (20-19=1). All this hints at the importanceof maintaining a sufficient bandwidth with minimal phase shift, while at the same time minimizing high frequency artifacts and distortions. All of the above and our experience listening and designing suggest that there are many subtle aspects to hearing that are beyond the realm of simple traditional measurement characterizations.The way in which an analog amplifier handles very small signals is as important as the way it behavesat high levels. For low distortion, an analog amplifier must have a linear transfer characteristic, in other words, the output signal must be an exact replica of the input signal, differing only in magnitude. The magnitude can be controlled by a gain control or fader (consisting of a high quality variable resistor that, by definition, has a linear transfer characteristic.) A dynamics controller - i.e. a compressor,limiter or expander - is a gain control that can adjust gain of the amplifier very rapidly in response to the fluctuating audio signal, ideally without introducing significant distortion, i.e. it must have a linear transfer characteristic. But, by definition, rapidly changing gain means that a signal “starting out” to be linear and, therefore without distortion, gets changed on the way to produce a different amplitude. Inevitably our data bank of “natural” sound is built up on the basis of our personal experience andthis must surely emphasize the importance of listening to “natural” sound, and high quality musical instruments within acoustic environments that is subjectively pleasing so as to develop keen awareness that will contribute to a reliable data bank. Humans who have not experienced enough “natural”5sound may well have a flawed data bank! Quality recording equipment should be capable of retaining “natural” sound and this is indeed the traditional measuring stick. And “creative” musical equipment should provide the tools to manipulate the sound to enhance the emotional appeal of the music without destroying it. Memory and knowledge of real acoustic and musical events may be the biggest tool and advantage any recording engineer may possess.One needs to be very careful when one hears traces of distortion prior to recording because some flavors of distortion that might seem acceptable (or even stylish) initially, may later prove to cause irreparable damage to parts of the sound (for example, “warm lows” but “harsh sibilance”) or in louder or quieter sections of the recording. Experience shows that mic preamps and basic console routing paths should offer supreme fidelity otherwise the engineer has little control or choice of recorded “color” and little recourse to undo after the fact. Devices or circuits that can easily be bypassed are usually better choices when “color” is a consideration and this particularly is an area where one might consider comparing several such devices. Beware that usually deviations from linearity carry at least as much long-term penalty as initial appeal, and that one should always be listening critically when recording and generally “playing it safe” when introducing effects that cannot be removed.1. Tsutomu Oohashi, Emi Nishina, Norie Kawai, Y oshitaka Fuwamoto, and Hishi Imai. NationalInstitute of Multimedia Education, Tokyo. “High Frequency Sound Above the Audible Range,Affects Brain Electric Activity and Sound Perception” Paper read at 91st. Convention of the A.E.S.October 1991. Section 7. (1), Conclusion.2. Miland Kunchur,Depart of Physics and Astronomy, University of South Carolina. “Temporal resolution of hearing probed by bandwidth restriction”, M. N. Kunchur, Acta Acustica united with Acustica 94, 594–603 (2008) (http://www.physics. /kunchur/Acoustics-papers.htm)3. Miland Kunchur,Depart of Physics and Astronomy, University of South Carolina.Probing the temporal resolution and bandwidth of human hearing , M. N. Kunchur, Proc. of Meetings on Acoustics (POMA) 2, 050006 (2008)517 USAGE NOTESThe 517’s feature set allows it to be used in many different ways in the studio. Here are a couple creative things to try:For vocals, take two mics, your favorite condenser plugged into the Mic Input, and a SM57 or SM58 set up 6 inches closer to the vocalist and plugged into another mic pre, then into the 517 Inst Input. Adjust Vari-Phase to taste, and maybe add some light compression from the 517 followed by your usual vocal compressor, which may behave even nicer because of that touch of “pre-compression” from the 517. The 517 can also be a very useful tool on multi-miced sources like drums. A snare drum mic for instance can be amplified, compressed and phase alligned to the rest of the kit (especially the overheads) when changing mic placement alone isn’t satisfying. This technique can be extremely useful when recording in a home studio environment, with fewer placement options available.Used with a synthesizer, the blend can be used to mix together or alternate between the tones from a room capture off an amp and the direct signal. As the blend is turned towards mic, the “room” signal becomes more prevalent. When blend is closer to direct, the signal is drier. The variphase, silk and compressor controls may also be incorporated for additional effects.6Amplified Instrument Use with Blended DI and Mic SignalsWhen used for instruments, the 517 can be used to phase align,combine and compress direct and miced instrument signals. Toachieve this, use the DI for the instrument’s direct signal and the micpreamp for the speaker cabinet signal. Connect the external ampli-fier to the passive DI thru on the 517 faceplate. The blend control isused for mixing direct and amplified signals to achieve the desiredtonality of the two sources, and the variphase is used to minimize orextenuate phase cancellations between the two signals. To compressthe blended signal, simply engage the compresor and adjust theconpressor threshold dial as desired. This technique could also beused to create a single, mixed output of a guitar and vocals, or anyother pairing of microphone and instrument signals.517 FEATURESMICROPHONE INPUTThe microphone input is balanced but not floating, being a variant of an instrumentation amplifier.Our well-proven “Transformer-Like-Amplifier” (T.L.A.) configuration is used, which includes an accurate toroidal Common Mode Low Pass Filter that rejects Common Mode signals and excludes frequencies above 150 kHz. (There are high powered broadcast transmitters at and above this frequency in several Continents and, even if you can’t hear them, any vestigial intermodulation products must be excluded!)When the Mic Gain switch is set to Unity (0 dB), the Portico 517 microphone pre-amplifier can handle a balanced input signal of more than +20 dBu without an input attenuator pad! This is a unique feature that enables this input to double as an additional line input.THE MIC AND INSTRUMENT / BLEND OUTPUTSThe main output signal comes from the output transformer secondary which is balanced and ground free. A ground free connection guarantees freedom from hum and radio frequency interference when connected to a balanced destination such as the input to another Portico module or a high quality ADC. However the transformer may be used with one leg grounded without any change in performance. Itis not necessary to “ground” one leg at the Portico output. It would normally get a ground connection when fed to equipment that is not balanced. Maximum output level is + 22 dBu, which provides a large margin over and above the likely maximum requirement of any destination equipment to which the 517 is connected.MIC GAINA 12-way precision rotary switch covering from Line (0) and Mic from 0 to 66 dB in 6 dB steps.DI GAINContinuous gain control from 0dB to 30dBIN and THRU 1/4” PHONE JACKSThese 2 jacks are used for DIRECT INJECTION (DI) or INSTRUMENT inputs and are simply paralleled and wired together. Inserting a plug into either jack breaks the normal MIC input and the user has the full range of MIC GAIN and TRIM. These jacks have a 3 mega ohm input impedance that will provide less loading (better highs) than most DI boxes, and the sheer amount of gain that is available makes these inputs extremely versatile.COMPRESSORWhen signals exceed the “threshold” level, the gain is reduced at a controlled 2:1 ratio, with attack and release time constants set to standard or fast depending on the internal jumper setting.SILKMuch could be written about this feature, suffice to say, that it gives a subtle option to enhance sound quality in the direction of vintage modules. The silk button reduces negative feedback and adjusts the frequency spectrum to provide a very sweet and musical performance. We suggest you try it and make 8your own judgment.BLENDCombines the Mic and DI signals on the DI / Blend Output.+48VBack panel switch makes phantom power available at the microphone input.POLARITYPush button inverts the polarity of the signal path. The symbol “Ø” is often used to denote opposite polarity.VARIPHASEThe VARPHASE control rotates the phase incrementally, allowing two signals with the same source to be phase aligned. This control is most apparent when the signals being combined are at roughly equal levels.INDICATORSIndicators on the 517 denote signal presence and clipping on both DI and mic signals, as well as compressor activation.INTERNAL JUMPERSTwo internal jumpers in the 517 can be set to change the compressor time constants and the engage the HPF.HIGH PASS FILTERThe high pass filter is a valuable aid in any signal chain but particularly so in a microphone preamplifier. Signals below 80hz can be attenuated at a rate of 12db / octave, getting rid of building rumble, air handling motor hum etc.INTERNAL JUMPERSThe compressor ships set to the fast time constant. To change the compressor time constant to slow, move the J1 jumper found under the top lid at the back of the unit, to the slow position.To engage the high pass filter, switch the J3 jumper position, found under the top lid at the back of the unit, to HPF in.SPECIFICATIONSFrequency Response:Mic PreMain Output, no load,-0.1dB @ 10Hz, -1 dB @ 120kHzDI InputMain Output, no load,-0.1dB @ 10Hz, -1 dB @ 120kHzNoise:Measured at Main Output, unweighted, 22Hz-22kHz,Terminated 150 Ohms.9With gain at unity better than –100 dBuWith gain at 66 dB better than –60.5 dBuEquivalent Input Noise better than –125 dBuDIWith Gain @ Unity Typically better than -100dBuWith Gain @ +30 dB Typically better than -75dBu High Pass Filter:Frequency: 80HzSlope: 12 dB/Octave BesselMicrophone Input Impedance10,000 OhmsInstrument Input Impedance1.6 Meg ohms (3.3 Meg Ohms balanced)Maximum Output Level:Maximum output from 20 Hz to 40 kHz is +23 dBu.Gain:Mic Pre:Unity to +66dB in 6dB stepsDI:Unity to +30dB continuous.T otal Harmonic Distortion and Noise:@ 1kHz, +20 dBu output: Main Output: Better than 0.001% @ 20Hz, +20 dBu output: Main Output: Better than 0.002% Silk Engaged: Better than 0.2% Second harmonicVariphase@100Hz: 2.9 degrees to 57.4 degrees.@1kHz: 33.5 degrees to 156 degrees.@10kHz: 149 degrees to -175 degrees.Crosstalk:Measured channel to channel: Better than –90 dB @ 15kHz. Phantom Power:+48 Volts DC +/- 1% (provided by 500 Series Rack)Power requirements:Supplied by 500 series rack with 110-125 mA @ +/- 16V CompressorThreshold: Continuously Variable from -20dBu to +10dBuRatio: Fixed at 2:110PRODUCT WARRANTYRupert Neve Designs warrants this product to be free from defects in materials and workmanship for a period of three (3)years from date of purchase, and agrees to remedy any defect identified within such three year period by, at our option, repairing or replacing the product.LIMITATIONS AND EXCLUSIONSThis warranty, and any other express or implied warranty, does not apply to any product which has been improperly installed, subjected to usage for which the product was not designed, misused or abused, damaged during shipping, damaged by any dry cell battery, or which has been altered or modified in any way. This warranty is extended to the original end user purchaser only. A purchase receipt or other satisfactory proof of date of original purchase is required before any warranty service will be performed. THIS EXPRESS, LIMITED WARRANTY IS IN LIEU OF ALL OTHER WARRANTIES, EXPRESS OR IMPLIED, TO THE EXTEND ALLOWED UNDER APPLICABLE STATE LAW. IN NO EVENT SHALL RUPERT NEVE DESIGNS BE LIABLE FOR ANY SPECIAL, INCIDENTAL, OR CONSEQUENTIAL DAMAGES RESULTING FROM THE USE OF THIS PRODUCT. Some states do not allow the exclusion or limitation of consequential damages or limitations on how long an implied warranty lasts, so this exclusion may not apply to you.WARRANTY SERVICEIf you suspect a defect in this product, please call us at 512-847-3013 or contact our support staff (service@rupertneve. com) for troubleshooting. If it is determined that the device is malfunctioning, we will issue a Return Material Authorization and provide instructions for shipping the device to our service department.Rupert Neve DesignsPO Box 1969Wimberley TX 78676tel: +1 512-847-3013fax: +1 512-847-8869775-00007 Rev D。

互调失真（Intermodulation distortion）Transient intermodulation distortion (referred to as TIM distortion), which was released publicly in 70s, is closely related to negative feedback. In the transient signal pulse input, the circuit capacitor sound output output voltage can not be due immediately, and the negative feedback circuit can not get timely response, AMP in open-loop state at this moment, the output moment overload causes clipping, the clipping distortion called transient intermodulation distortion, the distortion in the more serious on the amplifiers.Edit this paragraph principleAs everyone knows, negative feedback (Negative Feedback) is the role of the inverted phase output value is negative, then the feedback to the input and the set value of subtraction, the error signal, then the controller will be made according to the size of the error correction, thereby greatly reducing distortion. But due to the negative feedback to the input signal and feedback output signal subtraction, reduces the signal level, when the negative feedback quantity to the output signal and the input signal is reduced to the same level, namely the whole line without amplification, this amplifier is a buffer amplifier (Buffer Amplifier), it has the advantages of high input impedance, low output impedance that is often used for impedance matching. If we want to make a larger output signal level, the amplifier gain should be increased, which in the tubes and transistors machine is not difficult. But the negative feedback can effectively reduce distortion, but the distortion caused by the new transient intermodulation distortion, the distortion in the transistor (stone) machineis the most serious. This is because the stone used up to the depth of about 50-60dB negative feedback to improve the stability and reduce the distortion, although this machine will easily obtain technical parameters of transistor high. But there is gain and loss, in order to reduce the depth of negative high frequency parasitic oscillation caused by feedback, general stone to join a small capacitance between pre driver transistor collector and base frequency, the phase lag slightly, but no matter how the capacity of small, also want to have some time to charge, when the signal contains high speed transient pulse when the capacitor charging speed can not keep up, this moment is not in line with negative feedback, this time due to the input signal and feedback signal subtraction, causing the signal level is too strong, the amplifier circuit instantaneous overload (Overload) due to negative feedback, machine capacity, overload strength high, often up to more than a few times, then the output signal clipping occurs (Clipping) phenomenon, transient intermodulation distortion resulting in the distortion of the most because of stone, This distortion is therefore often referred to as a "transistor" sound.Edit this paragraph to reduce distortionThe transient intermodulation distortion is caused by the large gain in the delay time of the negative feedback amplifier. As long as the gain in this period is controlled, the transient intermodulation distortion can be eliminated. In order to eliminate the transient intermodulation distortion, we also add negative feedback signals at the negative input of the amplifier to control its gain while the signal enters the positive input of the amplifier. Although the negative feedbacktime delay is difficult to solve, but to reduce its impact, use large loop low negative feedback, so even if there is a negative feedback signal input time delay, but also strong; also available multistage negative feedback, such as feedback time, short path, not easy to induce transient intermodulation distortion. Besides, in the design should also try to use a variety of shielding and filtering measures to reduce the high frequency interference signal into the amplifier, the radio frequency interference while human ear can not hear, but their frequency is very high, easily induced transient intermodulation distortion. Transient intermodulation distortion is only when transient signals faster than the amplifier response ability range occurs, in addition, in addition to the distortion, another is ringing signal will fast (Ringing) distortion when the input signal speed and amplitude of the first hour, there is a ringing phenomenon, when the signal speed to a certain extent when transient intermodulation distortion will occur, but when the signal speed and amplitude, directly into the state of transient intermodulation distortion.All kinds of high speed but small amplitude of high frequency interference noise, most likely to cause ringing, this is the sound equipment to have a good anti-interference measures of a big reason.Edit transient intermodulation distortion of headphones in this sectionTransient intermodulation distortion refers to the transient signal pulse input, the circuit capacitor sound output outputvoltage can not be due immediately, and the negative feedback circuit can not get timely response, AMP in open-loop state at this moment, the output moment overload causes clipping, the clipping the distortion called transient intermodulation distortion, the distortion is more serious in the upper tube. Method of reducing this distortion are: transient intermodulation distortion is a dynamic index amp, amp mainly by the depth of the internal negative feedback caused by. Is arch-criminal affect the sound quality, resulting in "transistor sound tube" and "metal sound". Method of reducing this distortion are: 1. select devices and adjust work better, to improve the amplifier open-loop gain and open-loop frequency response. 2. strengthen the negative feedback of each amplifier stage and cancel the negative feedback of each loop. The influence of terminal devices on transient performance. Headphone amplifier is a circuit unit with ultra high magnification when applied to light reversal. It can be made up of discrete devices and can be implemented in semiconductor chips. With the development of semiconductor technology, most of the room is now in the form of a single. We often hear the amp conversion rate, strong signal conversion rate can be better and ensure timely signal amplification, and the smaller the conversion rate will make a sudden strong signal processing becomes delay. How much is the conversion rate is good, is not the focus of this paper, the relationship between the parameters and understand some transient enough.Edit this paragraph in detailTransient Intermodulation Distortion, also known as TIM distortion, is the culprit of transistor noise, transientintermodulation distortion. The TIM method was not published until 70s. Since transient intermodulation distortion is closely related to negative feedback, it is necessary to start with negative feedback when discussing transient intermodulation distortion. Negative feedback (Negative Feedback) is a widely used in the field of engineering technology, control technology is simple and practical, is a negative feedback control loop control technology (Close Loop Control) is a part of the system, but because of a wide range of applications, so often used as a synonym for closed loop control. Negative feedback is actually a kind of natural law, widely exists in people's daily life for example, when we are driving a car, if the car was found to deviate from the route, we will be in the opposite direction twisting the steering wheel, the car back to the correct line. Here is our eyes as a negative feedback path, is responsible for the output value (car driving direction) back to the mining device (brain), then the controller output value and set value (the right direction) compared to each other (subtraction), then according to the error comparison, a correction signal (twisted wheel) to correct. Thus, the negative feedback is the role of the output phase (becomes negative), then the feedback to the input terminal, and set the value of subtraction, the error signal, then the controller will be made according to the size of the error correction. In the electronic amplifying circuit, due to changes in parts of the nonlinear, symmetry, temperature, noise and other reasons, so that the signal was amplified at the same time, no distortion can avoid being added all sorts of negative feedback can effectively reduce the distortion. For a simple example, if the amplifier amplifies a sinusoidal signal, the output is distorted because of the nonlinearity, symmetry, andtemperature of the element. The distorted signal is compared with the input signal by negative feedback to reduce distortion. Because the output and input subtraction, although the stability of the gain, but also a substantial increase in the amount of reduction. If the output signal is amplified to a sufficient intensity,Amplifier amplification (gain) will increase, but fortunately, this is not difficult, especially transistors. If we will increase the amount of negative feedback, the output signal is reduced to the same level and the input signal level, i.e. no amplification, the amplifier circuit has a special name, called a buffer amplifier (Buffer Amplifier). Although the signal has not been amplified, the amplifier has high input impedance and low output impedance. So buffer amplifiers are often used as impedance matching. Since negative feedback can reduce distortion effectively, why does it cause transient intermodulation distortion? The original problem lies in the time, in which the transistor machine is the most serious. Compared to the vacuum tube, transistor has durable, small volume, light weight and higher rate of advantages, its disadvantages are characteristic of unstable, easily affected by factors such as temperature and distortion and even out of control. One solution is to use deep negative feedback of up to 50 to 60dB. Transistors, however, have high amplification rates, sacrificing some of them. Because of the large depth of negative feedback, a significant reduction in distortion is made, so transistor machines can easily acquire superb specifications. But the trouble is it, in order to reduce the depth of negative high frequency parasitic oscillation caused by feedback, transistor amplifier general to join a smallcapacitor between the front driving transistor base and collector, the phase frequency slightly lag, known as the lag price or according to the premium, but no matter how small capacitance in general, it will take time to charge, when the input pulse signals with transient high speed, small capacitance to charge, that is to say at this moment is not in line with negative feedback state. Since the input signal is not subtracted from the negative return signal and the signal is too strong, these excessive signals will cause the instantaneous overload of the amplification line (Overload). Because transistors have large amount of negative feedback and higher signal intensity, they often reach tens or even hundreds of times, resulting in the output signal Xiao Bo (Clipping). This is transient intermodulation distortion, because at most the transistor circuit occurs, so it is also known as "atomic" or "transistor" sound. This negative feedback time delay problem is also frequently encountered in industrial control systems, known as the Time (Dead) problem, mostly because the Sensor is too far away. For example, in a constant temperature water heater, the temperature probe is installed in the position away from the heating line. As a result, when the detector senses the water temperature is sufficient, the water temperature near the heater has already overheated. The result of this control must be that the water temperature swings sharply between overheating and supercooling, referred to as controlled overshoot (Overshoot) or system oscillation. The pure delay is still a big problem that puzzles the automatic control technology, and the papers about the solution are less than 1000 since 50s. But there is no simple and effective way to solve them. Although the negative feedback time delay is not a good deal, but also have no way to solve, we can not simplymake it appear, or even its not to cause too much damage, there are a variety of methods, such as only small amounts of large loop negative feedback, so named negative feedback time delay, the input signal is not too strong. The reduced negative feedback is replaced by partial negative feedback across 1 amplification stages. The local negative feedback path is short and time fast, and it is difficult to induce transient intermodulation distortion. The vacuum tube is stable, not necessarily with deep negative feedback inhibition of distortion, and distortion is the most even order harmonic distortion to the human ear so Danji no general so-called "atom". As for other methods used to prevent transient intermodulation distortion in line design, because of the more boring theories involved, it is not introduced here. In addition to the circuit design to prevent transient intermodulation distortion, fancier can also take another measure to reduce the transient intermodulation distortion, that is to use a variety of shielding and filtering measures to reduce high frequency interference signal into the amplifier,Although many of these signals are invisible to human ears, radio frequency interference is very easy to induce transient intermodulation distortion, so that the input stage is overloaded and the music signal can not be amplified normally. The setting of negative feedback has a great influence on the performance of PA. General power amplifier circuit, negative feedback from the output. The distortion produced by the current amplifier stage is improved by the large loop negative feedback. This kind of feedback often makes the power amplifier objective distortion index improved, while subjective listening is not satisfactory. The final distortion generatedby negative feedback input level, through the pre amplification of the compensation and the compensation adjustment, and adjustment is necessarily slow, is bound to make the system transient response speed reduced, easy to induce transient intermodulation distortion (TIM), and the high frequency signal distortion and phase shift, in the sense of hearing for transistor sound blunt "". In addition, the back EMF generated by the loudspeaker and the radio frequency interference induced by the speaker line are polluted by the signal, which affects the purity of the sound quality. In order to avoid the above disadvantages, the front voltage feedback can be used and the capacitor is used to isolate the front stage from the final stage. Since this capacitor is located on the signal channel, the metallized polypropylene capacitor is used to ensure the pure sound quality. Thus, the final stage becomes a 0dB stage amplifier without negative feedback (pure current amplifier), so the preamplifier is taken as a high gain amplifier at this stage. Many methods are in practice, and there is not yet a generally easy way to solve the transistor sound".。

ISSN 1002-4956 CN11-2034/T实验技术与管理Experimental Technology and Management第38卷第2期2021年2月Vol.38 No.2 Feb. 2021D O I: 10.16791/ki.sjg.2021.02.044基于C ST仿真软件的阻抗匹配设计教学实验赓臻\賡志斌2，刘宇平2(1.杭州电子科技大学电子信息学院，浙江杭州310018;2.新余学院数学与计算机学院，江西新余338000 )摘要：传输线的阻抗匹配是电磁场与微波技术中一个重要的理论，是射频微波电路设计的基础:但相关概念较为抽象，传统教学过程以数学推导为主，学生理解困难。

为了增强学生对阻抗匹配的理解，以微带线阻抗匹配的典型工程应用为案例，将理论分析与电磁仿真相结合，对微带线阻抗匹配网络进行设计，增强学生对传输线阻抗匹配的理解:使学生从理论到仿真，从数学推导到可视化的验证，构建全面的知识体系，增强 解决复杂工程问题的能力。

关键词：阻抗匹配；单支节匹配网络；微带线；电磁仿真中图分类号：G433 文献标识码：A 文章编号：1002-4956(2021)02-0204-04Teaching experiment of impedance matching designbased on CST simulation softwareLIAO Zhen1,LIAO Zhibin2,LIU Yuping2(1. School of Electronics and Information, Hangzhou Dianzi University, Hangzhou 310018, China;2. School of Mathematics and Computer, Xinyu University, Xinyu 338000, China)A bstract: The theory o f transmission line impedance matching is an important theory in electromagnetic field and microwave technology, and it is the fundamental o f radio and microwave circuit design. But the relative concepts are abstract and teaching process is based on mathematical derivation, which makes it difficult for students to understand. By taking a typical project o f the microstrip impedance matching as an example, the impedance matching network is designed by combining theoretical deduction with simulation, which has enhanced students’understanding o f transmission line impedance. The experiment is helpful to construct a comprehensive knowledge structure from theory to simulation and from formula deprivation to visual presentation and enhance students1 ability to solve complex engineering problems.Key w ords: impedance matching; single-stub matching network; microstrip; electromagnetic simulation随着通信技术的蓬勃发展，社会对射频微波技术 人才的需求也与日俱增+3]。

6.1.1 真空磁导率(permeability of vacuum)6.1.2 介电系数，电容率(permittivity)0也称电常数(electric constant)。

6.1.3电动势(electromotive force)6.1.4接触电动势(contact electromotive force)6.1.5感应电(动)势(induced electromotive force)6.1.6导体(conductor)6.1.7绝缘体(insulator)6.1.8半导体(semiconductor)6.1.9超导体(superconductor)6.1.10接触电位(差)（contact potential[difference]）6.1.11热电效应(thermoelectric effect)6.1.12塞贝克效应(Seebeek effect)6.1.13珀耳帖效应(Polfier effect)6.1.14汤姆逊效应(Thomson effect)6.1.15约瑟夫森效应(Josephson effect)6.1.16量子化霍尔效应(quantum Hall effect)6.1.17单电子隧道效应(single electron tunnel effect)6.1.18 功率天平（Watt balanc e）6.1.19交流电阻时间常数(time constant of ac resistor)6.1.20介电强度(dielectric strength)6.1.21绝缘电阻(insulation resistance)6.1.22 电流(electric current)6.1.23 电压(voltage)6.1.24 电阻(resistance)6.1.25 电导(conductance)6.1.26 阻抗(impedance)6.1.27 导纳(admittance)6.1.28 电容(capacitance)6.1.29 电感(inductance)6.1.30 电阻率(resistivity)6.1.31 电导率(conductivity)6.1.32 磁导率（permeability）6.1.33 静电场(electrostatic field)6.1.34 电场强度(electric field intensi ty) 6.1.35 电位(electric potential)6.1.36 电荷(electric charge)6.1.37 库伦定律(Coulomb’s law)6.1.38 电位移(electric displacement)6.1.39 拉普拉斯方程(Laplace’s equation) 6.1.40 静电感应(electrostatic induction) 6.1.41 恒定电场(steady electric fiel d) 6.1.42 欧姆定律(Ohm l aw)6.1.43 焦耳定律(Joule’s l aw)6.1.44 安培(ampere)6.1.45 伏特(volt)6.1.46 库仑(couomb)6.1.47 欧姆(ohm)6.1.48 西门子(siemens)6.1.49 法拉(farad)6.1.50 亨利(henry)6.1.51 瓦特(watt)6.1.52 电路(electric circuit)6.1.53 激励(excitation)6.1.54 响应(response)6.1.55 电路元件(electric circuit elements)6.1.56 无源二端元件(passive two-terminal elements)6.1.57 电压源(voltage sources)6.1.58 电流源(current sources)6.1.59 受控源(controlled sources)6.1.60 开路(open circuit)6.1.61 短路(short circui t)6.1.62 理想变压器(ideal transformer)6.1.63 基尔霍夫定律(Kirchhoff’s law)6.1.64 直流(direct current)6.1.65 交流(alternating current)6.1.66 正弦电流(sinusoidal current)6.1.67 频率(frequency)6.1.68 赫兹(hertz)6.1.69 相位(phase)6.1.70 相量(phasor)6.1.71相量图(phasor diagram)6.1.72 谐振(resonance)6.1.73 铁磁谐振电路(ferro- resonance circuit)6.1.74 三相电路(three-phase circuit)6.1.75 三相电源(three-phase sources)6.1.76 三相负载(three-phase loads)6.1.77 相电压(phase voltages)6.1.78 线电压(line voltages)6.1.79 相电流(phase currents)6.1.80 线电流(line currents)6.1.81 对称三相电路(symmetrical three-phase circuit)6.1.82 非对称三相电路(unsymmetrical three-phase circuit)6.1.83 三相电路功率(power of three-phase circuit)6.1.84 非正弦周期电流电路(non-sinusoidal periodic current circuits)6.1.85 基波电流(fundamental current)6.1.86 谐波电流(harmonic current)6.1.87 频谱(frequency spectrum)6.1.88 瞬时值(instantaneous value)6.1.89 平均值(average value)6.1.90有效值(effective value)6.1.91 峰值（peak [value]）6.1.92波形因数(wave factor)6.1.93 总谐波畸变率(total harmonic distortion)6.1.94 平均功率(average power)6.1.95视在功率(apparent power)6.1.96无功功率(reactive power)6.1.97 复功率(complex power)6.1.98 谐波功率(harmonic power)6.1.99 畸变功率(distortion power)6.1.100 伏安(volt ampere)6.1.101 乏(var)6.1.102 瓦特小时(watt hour)6.1.103 串联(series connection)6.1.104 并联(parallel connection)6.1.105 星形阻抗与三角形阻抗的变换(transformation between star-connected and delta connected impedances)6.1.106电源的等效变换(equivalent transformation between sources)6.1.107回路法(loop analysis)6.1.108节点法(node analysi s)6.1.109叠加定理(superposition theorem)6.1.110替代定理(substitution theorem)6.1.111 互易定理(reciprocity theorem)6.1.112戴维南定理(Thevenin theorem)6.1.113诺顿定理(Norton theorem)6.1.114 二端口(2-port)6.1.115 特性阻抗(characteristic impedance)6.1.116 输入阻抗(input impedance)6.1.117 输出阻抗(output impedance)6.1.118 传播常数(propagation constant)6.1.119 品质因数(quality factor )6.1.120 阻抗匹配(impedance matching)6.1.121 网络函数(network functions)6.1.123 分布参数电路(distributed parameter circuit)6.1.124 一阶电路(first order circuit)6.1.125 二阶电路(second order circuit)6.1.126 高阶电路(high order circui t)6.1.127 非线性电路(nonlinear electric circuit)6.1.128 端子（terminal）6.1.129 端变量（terminal variable）6.1.130 两端(2T) (2-terminal)6.1.131 三端(3T) (3-terminal)6.1.132 四端(4T) (4-terminal)6.1.133五端(5T) (5-terminal)6.1.134四端对(4TP)(4-terminal pair)6.1.135磁场(magnetic fiel d)6.1.136 磁感应强度(magnetic induction)6.1.137磁通量(magnetic flux)6.1.138 磁导率(permeability)6.1.139 相对磁导率(Reletive permeability) 6.1.140磁矩（Magnetic(area) moment）6.1.141 磁化强度（Magnetizat ion）6.1.142 磁极化强度(magnetic polarization) 6.1.143 磁场强度（magnetic intensity）6.1.144磁偶极矩（magnetic dipole moment）6.1.145 磁通势（magnetomotive force）6.1.146 磁阻（reluc tanc e）6.1.147 磁导（permeanc e）6.1.148 磁化率（magnetic susceptibility）6.1.149 磁共振（magnetic resonance）6.1.150核磁共振（nuclear magnetic resonance）6.1.151霍尔效应（hall effect）6.1.152 波尔磁子（Bohr magneton）6.1.153 质子旋磁比（Proton gyro magnetic ratio）6.1.154 磁通量子（F1ux quantum (F1uxon)）6.2 电学计量6.2.1.1直流电压基准(Primary Standard of DC V oltage)6.2.1.2直流电动势基准(Primary Standard of DC Electromotive Force)6.2.1.3直流电阻基准(Primary Standard of DC Resistance)6.2.1.4电容基准(Primary Standard of Capacitance)6.2.1.5电容器损耗因数基准(Primary Standard of Dissipation Factor)6.2.1.6电感基准(Primary Standard of Inductance)6.2.1.7交流电流基准(Primary Standard of AC Current)6.2.1.8交流电压基准(Primary Standard of AC V oltage)6.2.1.9交流功率基准(Primary Standard of AC Power)6.2.1.10工频电能基准(Primary Standard of AC Energy at Industrial Frequency)6.2.1.11磁感应强度基准(Primary Standard of Magnetic Flux Density)6.2.1.12数字阻抗电桥标准(Standard for LCR meter)6.2.1.13数字多用表检定装置(Standard of Multimeter)6.2.1.14超导强磁场标准(Standard of Supper Conducting High Magnetic Fi el d)6.2.1.15非铁磁金属电导率标准(Standard of Conductivity for Nonferrous Metals)6.2.1.16模/数、数/模转换测量标准（Standard of ADC and DAC）6.2.1.17标准电池(standard cell)6.2.1.18固态电压标准(solid state voltage standard)6.2.1.19标准电阻(standard resistor)6.2.1.20计算电容(cross capacitor)6.2.1.21感应分压器(inductive voltage divider)6.2.1.22分流器(shunt)6.2.1.23直流电流比较仪(direct current comparator)6.2.1.25多功能校准源(multifunction calibrator)6.2.1.26数字阻抗电桥(LCR meter)6.2.1.27 电压表（voltmeter）6.2.1.28 电流表(amperometer)6.2.1.29 电阻表(ohnneter)6.2.1.30 功率表(Watt meter)6.2.1.31 电能表(kWh meter)6.2.2电学计量常用测量方法6.2.2.1 直接测量（法）(direct (method of) measurement)6.2.2.3组合测量（法）(combination (method of) measurement)6.2.2.4 比较测量（法）(comparison (method of) measurement)6.2.2.5 零值测量（法）(null (method of) measurement)6.2.2.6 差值测量（法）(differential (method of) measurement)6.2.2.7 替代测量（法）(substitution (method of) measurement)6.2.2.8 不完全替代法(semi-substitution method of measurement)6.2.2.9 内插测量（法）(interpolation (method of) measurement)6.2.2.10 互补测量（法）(complementary (method of) measurement)6.2.2.11 差拍测量（法）(beat (method of) measurement)6.2.2.12 谐振测量（法）(resonance (method of) measurement)6.2.2.13 模数转换(analogue to digital conversion)6.2.2.14 数模转换(digital to analogue conversion)6.2.2.15 静电屏蔽(electrostatic screen)6.2.2.16 磁屏蔽(magnetic screen)6.2.2.17 泄漏电流(leakage current)6.2.2.18 电位屏蔽(potential screen)6.2.2.19 等电位屏蔽(equip—potential screen)6.2.2.20 无定向结构(astatic construction)6.2.2.21交流-直流转换(AC-DC conversion)6.2.2.22交流-直流转换器<AC-DC converter)6.2.2.23交流-直流比较仪(AC-DC comparator)6.2.2.24热电变换器(thermal converter)6.2.2.25 共模电压(common mode voltage)6.2.2.26 串模电压(series mode voltage)6.2.2.27 共模抑制比(common mode rejection ratio ———CMRR)6.2.2.28 串模抑制比(series mode rejection ratio——SMRR)6.2.2.29 非对称输入(asymmetrical input)6.2.2.30 非对称输出(asymmetrical output)6.2.2.31 对称输入(symmetrical input)6.2.2.32 对称输出(symmetrical output)6.2.2.33 差分输入电路(differential input circuit)6.2.2.34 接地输入电路(earthed input circuit 或grounded input)6.2.2.35 接地输出电路(earthed output circuit或grounded output)6.2.2.36 浮置输入电路(floating input circuit)6.2.2.37 浮置输出电路(floating output circuit)6.2.3.1 模拟（测量）仪表（analogue (measuring) instrument）模拟指示仪表（analogue indicating instrument）6.2.3.2 数字（测量）仪表（digital (measuring) instrument）6.2.3.3 热电系仪表（electrothermal instrument）6.2.3.4 双金属系仪表（bimetallic instrument）6.2.3.5 热偶式仪表（thermocouple instrument）6.2.3.6 整流式仪表（rectifier instrument）6.2.3.7 振簧系仪表（vibrating reed instrument）6.2.3.8 多用表、万用表（multimeter）6.2.3.9（测量）电桥（(measuring) bridge）6.2.3.10（测量）电位差计（(measuring) potentiometer）6.2.3.11 分压器（voltage divider）6.2.3.12 比较仪（comparator）6.2.3.13 指针式仪表（pointer instrument）6.2.3.14 光标式仪表（instrument with optical index）6.2.3.15 动标度仪表（moving-scale instrument）6.2.3.16 影条式仪表（shadow column instrument）6.2.3.17 静电系仪表（electrostatic instrument）6.2.3.18 磁电系仪表（(permanent magnet) moving-coil instrument）6.2.3.19 动磁系仪表（moving magnet instrument）6.2.3.20 电磁系仪表（moving-iron instrument）6.2.3.21 电动系仪表（electrodynamic instrument）6.2.3.22 铁磁电动系仪表（ferrodynamic instrument）6.2.3.23 感应系仪表（induction instrument）。

电气专业工程电气术语英文对照表电路，electric circuit电气工程electrical engineering电机electric machine自然科学physical science电气设备 electrical device电器元件 electrical element正电荷positive charge负电荷negative charge直流direct current交流alternating current电压voltage导体conductor功work电动势electromotiveforce电势差potential difference功率power极性polarity能量守恒定律the law of conservation energy变量variable电阻 resistance电阻率resistivity绝缘体insulator电阻器resistor无源元件passive element常数constant电导conductance短路short circuit开路open circuit线性的linear串联series并联parallel电压降voltage drop等效电阻equivalent resistance电容器capacitor电感器inductor储能元件storage element电场electric field充电 charge放电discharge动态的dynamic电介质dielectric电容capacitance磁场magnetic field电源power supplu变压器transformer电机electric motor线圈coil电感inductance导线conducting wire绕组wingding漏电阻leakage resistance电子系统electronic system结构图block diagram功能模块functional block放大器amplifier滤波器filter整形电路wave-shaping circuit振荡器oscillator增益gain输入阻抗input impedance带宽bandwidth晶体管transistor集成电路integrated circuit电力电子power electronics数字信号处理digital signal-processing 输出装置output device模拟信号analog signal数字信号digital signal传感器transducer采样值sample value模数转换器analog-to-digital converter 频谱frequency content采样频率sampling rate or frequendy 扰动disturbance分立电路discrete circuit数字化信号digitized signal运算放大器operational amplifier有源电路active circuit电子部件electronic unit封装package管脚pin同相端noninverting terminal反相输入inverting input电路图circuit diagram压控电压源voltage-controlled voltage source 开环增益open-loop gain闭环增益closed-loop gain负反馈negative feedback正饱和positive saturation线性区linear region电压跟随器voltage follower等效阻抗equivalent impedance逻辑变量logic variable位bit数字字digital word字节byte半字节nibble与运算AND operation真值表truth table与门AND gate非门NOT gate或门OR gate加号addition sign与非门NANA gate异或运算XOR operation逻辑表达式logic expression二进制binary system正逻辑positive logic负逻辑negative logic参考方向reference direction理想变压器ideal transformer电气绝缘electrical isolation阻抗匹配impedance matching电力electrical pewer绝缘变压器isolating transformer电压互感器voltage transformer电流互感器current transformer原边绕组primary winding工作频率operating frequency配电变压器distribution transformer电力变压器power transformer磁通密度flux density磁场magnetic field铁芯变压器iron-core transformer大功率high-power空芯air-core磁耦合magnetic coupling 小功率lower-power励磁损耗magnetizing loss磁滞损耗hysteresis loss涡流eddy current励磁电流exciting current漏磁通leakage flux互磁通 mutual flux线圈coil芯式core form壳式shell form高压绕组high-voltage winding磁链flux linkage电动势electromotive force有效值root mean square value匝数比turns ratio视在功率apparent power匝数the number of turns升压变压器step-up transformer 降压变压器step-down transformer电动机motor发电机generator机械能mechanical energy电能electrical energy电磁的electromagnetic直线式电动机linear motor同步电机synchronous machine感应电机induction machine定子stator转子rotor气隙air gap轴shaft电枢armature励磁绕组field winding无功功率reactive power制动状态braking mode稳态steady-state相序phase sequence反响制动plugging滞后电流lagging current励磁电抗magnetizing reactance 启动电流starting current变频器frequency changer感应电势induced voltage逆变器inverter周波变换器cycloconverter换向器commutator自动控制automatic control 控制器controller扰动disturbance期望值desired value压力pressure液位liquid level被控变量controlled variable 方框图block diagram传递函数transfer function工程控制process control伺服系统servomechanism流率flow rate加速度acceleration前向通路forward path补偿correction反馈通路feedback path闭环closed-loop开环open-loop输出output增益gain手动调节manual adjustment 变送器transducer误差error控制方式control mode比例控制proportional control 积分控制integral control微分控制derivative control执行元件manipulating element 调节时间setting time残差residual error不确定度uncertainty观测数据observations采样sample算术平均arithmetic average期望值expected value标准偏差standard deviation下限lower range limit上限upper range limit跨度span分辨率resolution死区dead band灵敏度sensitivity阈值threshold可靠性reliability过量程overrange恢复时间recovery time过载overload过量程极限overrange limit漂移drift准确性accuracy误差error重复性repeatability系统误差systemic error再现性reproducibility校准calibration线速度linear velocity角速度angular velocity弧度radian测速仪tachometer增量式编码器incremental encoder定时计数器timed counter稳定性stability接口interface调节器conditioner开关switch执行器actuator电磁阀solenoid valve连续控制系统sequential control system 触点contact常开normally open常闭normally closed限位开关limit switch继电器relay延时继电器time-delay relay接通电流pull-in current开断电流drop-out current电机启动器motor starter接触器contactor自锁触点holding contact整流器rectifier变流器converter逆变器inverter二极管diode阳极anode阴极cathode正向偏置forward biased反向偏置reverse biased阻断block稳压二极管zener diode晶体管transistor集电极collector基极base发射极emitter共发射极common-emitter双向晶闸管triac正半周positive half-cycle触发电流trigger circuit功率容量power capability功率器件power device晶闸管thyristor导通conduction正向阻断 forward-blocking通态on-state关断状态off-state反向击穿电压reverse breakdown voltage 漏电流leakage current电流额定值current rating漏极drain门极gate缓冲电路snubber circuit均流current sharing额定电压rated voltage可控开关controllable switch相控phase-controlled充电器charger工频line-frequency变换器converter整流rectification逆变inversion可逆调速revesible-speed再生制动regenerative barking关断时间turn-off time纯电阻负载pure resistive load脉动ripple感性负载inductance load周期time period带内部直流电动势的负载load witn an internal DC voltage 波形waveform换相commutation稳态steady state交流侧AC-side延时角delay angle交点intersection电力系统power system发电厂generating plant发电机generator负荷load输电网transmission nerwork配电网distribution network电electricity天然气natural gas原理图schematic diagram锅炉boiler热效率thermal efficiency风力wind power断路器circuit breaker变电所substation故障fault过电压overvoltage击穿值breakdown value过电流over current可靠性reliability继电器relay触点contact电流互感器current transformer合闸线圈operating coil分闸线圈trip coilCircuit theory is also valuable to students specializing in other branches of the physical science because circuit are a good model for the study of energy system in general,and because of the applied mathematics,physics,and topology involved.电路理论对于专门研究自然科学其他分支的学生来说也十分有价值，因为电路一般可以很好地作为能量系统研究的模型，并且电路理论涉及应用数学、物理学和拓扑学的相关知识。

英文名称：impedance matching基本概念信号传输过程中负载阻抗和信源内阻抗之间的特定配合关系。

一件器材的输出阻抗和所连接的负载阻抗之间所应满足的某种关系，以免接上负载后对器材本身的工作状态产生明显的影响。

对电子设备互连来说，例如信号源连放大器，前级连后级，只要后一级的输入阻抗大于前一级的输出阻抗5-10倍以上，就可认为阻抗匹配良好；对于放大器连接音箱来说，电子管机应选用与其输出端标称阻抗相等或接近的音箱，而晶体管放大器则无此限制，可以接任何阻抗的音箱。

匹配条件①负载阻抗等于信源内阻抗，即它们的模与辐角分别相等，这时在负载阻抗上可以得到无失真的电压传输。

②负载阻抗等于信源内阻抗的共轭值，即它们的模相等而辐角之和为零。

这时在负载阻抗上可以得到最大功率。

这种匹配条件称为共轭匹配。

如果信源内阻抗和负载阻抗均为纯阻性，则两种匹配条件是等同的。

阻抗匹配是指负载阻抗与激励源内部阻抗互相适配，得到最大功率输出的一种工作状态。

对于不同特性的电路，匹配条件是不一样的。

在纯电阻电路中，当负载电阻等于激励源内阻时，则输出功率为最大，这种工作状态称为匹配，否则称为失配。

当激励源内阻抗和负载阻抗含有电抗成份时，为使负载得到最大功率，负载阻抗与内阻必须满足共扼关系，即电阻成份相等，电抗成份绝对值相等而符号相反。

这种匹配条件称为共扼匹配。

阻抗匹配（Impedance matching）是微波电子学里的一部分，主要用于传输线上，来达至所有高频的微波信号皆能传至负载点的目的，不会有信号反射回来源点，从而提升能源效益。

史密夫图表上。

电容或电感与负载串联起来，即可增加或减少负载的阻抗值，在图表上的点会沿著代表实数电阻的圆圈走动。

如果把电容或电感接地，首先图表上的点会以图中心旋转180度，然后才沿电阻圈走动，再沿中心旋转180度。

重覆以上方法直至电阻值变成1，即可直接把阻抗力变为零完成匹配。

共轭匹配在信号源给定的情况下，输出功率取决于负载电阻与信号源内阻之比K，当两者相等，即K=1时，输出功率最大。

6.1.1 真空磁导率(permeability of vacuum) 6.1.2 介电系数，电容率(permittivity)0也称电常数(electric constant)。

6.1.3电动势(electromotive force)6.1.4接触电动势(contact electromotive force) 6.1.5感应电(动)势(induced electromotive force)6.1.6导体(conductor)6.1.7绝缘体(insulator)6.1.8半导体(semiconductor)6.1.9超导体(superconductor)6.1.10接触电位(差)（contact potential[difference]）6.1.11热电效应(thermoelectric effect)6.1.12塞贝克效应(Seebeek effect)6.1.13珀耳帖效应(Polfier effect)6.1.14汤姆逊效应(Thomson effect)6.1.15约瑟夫森效应(Josephson effect)6.1.16量子化霍尔效应(quantum Hall effect) 6.1.17单电子隧道效应(single electron tunnel effect)6.1.18 功率天平（Watt balance）6.1.19交流电阻时间常数(time constant of ac resistor)6.1.20介电强度(dielectric strength)6.1.21绝缘电阻(insulation resistance)6.1.22 电流(electric current)6.1.23 电压(voltage)6.1.24 电阻(resistance)6.1.25 电导(conductance)6.1.26 阻抗(impedance)6.1.27 导纳(admittance)6.1.28 电容(capacitance)6.1.29 电感(inductance)6.1.30 电阻率(resistivity)6.1.31 电导率(conductivity)6.1.32 磁导率（permeability）6.1.33 静电场(electrostatic field)6.1.34 电场强度(electric field intensity)6.1.35 电位(electric potential)6.1.36 电荷(electric charge)6.1.37 库伦定律(Coulomb’s law)6.1.38 电位移(electric displacement)6.1.39 拉普拉斯方程(Laplace’s equation) 6.1.40 静电感应(electrostatic induction)6.1.41 恒定电场(steady electric field)6.1.42 欧姆定律(Ohm law)6.1.43 焦耳定律(Joule’s law)6.1.44 安培(ampere)6.1.45 伏特(volt)6.1.46 库仑(couomb)6.1.47 欧姆(ohm)6.1.48 西门子(siemens)6.1.49 法拉(farad)6.1.50 亨利(henry)6.1.51 瓦特(watt)6.1.52 电路(electric circuit)6.1.53 激励(excitation)6.1.54 响应(response)6.1.55 电路元件(electric circuit elements) 6.1.56 无源二端元件(passive two-terminal elements)6.1.57 电压源(voltage sources)6.1.58 电流源(current sources)6.1.59 受控源(controlled sources)6.1.60 开路(open circuit)6.1.61 短路(short circuit)6.1.62 理想变压器(ideal transformer)6.1.63 基尔霍夫定律(Kirchhoff’s law)6.1.64 直流(direct current)6.1.65 交流(alternating current)6.1.66 正弦电流(sinusoidal current)6.1.67 频率(frequency)6.1.68 赫兹(hertz)6.1.69 相位(phase)6.1.70 相量(phasor)6.1.71相量图(phasor diagram)6.1.72 谐振(resonance)6.1.73 铁磁谐振电路(ferro- resonance circuit)6.1.74 三相电路(three-phase circuit)6.1.75 三相电源(three-phase sources)6.1.76 三相负载(three-phase loads)6.1.77 相电压(phase voltages)6.1.78 线电压(line voltages)6.1.79 相电流(phase currents)6.1.80 线电流(line currents)6.1.81 对称三相电路(symmetricalthree-phase circuit)6.1.82 非对称三相电路(unsymmetrical three-phase circuit)6.1.83 三相电路功率(power of three-phase circuit)6.1.84 非正弦周期电流电路(non-sinusoidal periodic current circuits)6.1.85 基波电流(fundamental current)6.1.86 谐波电流(harmonic current)6.1.87 频谱(frequency spectrum)6.1.88 瞬时值(instantaneous value)6.1.89 平均值(average value)6.1.90有效值(effective value)6.1.91 峰值（peak [value]）6.1.92波形因数(wave factor)6.1.93 总谐波畸变率(total harmonic distortion)6.1.94 平均功率(average power)6.1.95视在功率(apparent power)6.1.96无功功率(reactive power)6.1.97 复功率(complex power)6.1.98 谐波功率(harmonic power)6.1.99 畸变功率(distortion power)6.1.100 伏安(volt ampere)6.1.101 乏(var)6.1.102 瓦特小时(watt hour)6.1.103 串联(series connection)6.1.104 并联(parallel connection)6.1.105 星形阻抗与三角形阻抗的变换(transformation between star-connected and delta connected impedances)6.1.106电源的等效变换(equivalent transformation between sources)6.1.107回路法(loop analysis)6.1.108节点法(node analysis)6.1.109叠加定理(superposition theorem)6.1.110替代定理(substitution theorem)6.1.111 互易定理(reciprocity theorem)6.1.112戴维南定理(Thevenin theorem)6.1.113诺顿定理(Norton theorem)6.1.114 二端口(2-port)6.1.115 特性阻抗(characteristic impedance) 6.1.116 输入阻抗(input impedance)6.1.117 输出阻抗(output impedance) 6.1.118 传播常数(propagation constant)6.1.119 品质因数(quality factor )6.1.120 阻抗匹配(impedance matching)6.1.121 网络函数(network functions)6.1.123 分布参数电路(distributed parameter circuit)6.1.124 一阶电路(first order circuit)6.1.125 二阶电路(second order circuit)6.1.126 高阶电路(high order circuit)6.1.127 非线性电路(nonlinear electric circuit)6.1.128 端子（terminal）6.1.129 端变量（terminal variable）6.1.130 两端(2T) (2-terminal)6.1.131 三端(3T) (3-terminal)6.1.132 四端(4T) (4-terminal)6.1.133五端(5T) (5-terminal)6.1.134四端对(4TP)(4-terminal pair)6.1.135磁场(magnetic field)6.1.136 磁感应强度(magnetic induction)6.1.137磁通量(magnetic flux)6.1.138 磁导率(permeability)6.1.139 相对磁导率(Reletive permeability) 6.1.140磁矩（Magnetic(area) moment）6.1.141 磁化强度（Magnetization）6.1.142 磁极化强度(magnetic polarization) 6.1.143 磁场强度（magnetic intensity）6.1.144磁偶极矩（magnetic dipole moment）6.1.145 磁通势（magnetomotive force）6.1.146 磁阻（reluctance）6.1.147 磁导（permeance）6.1.148 磁化率（magnetic susceptibility）6.1.149 磁共振（magnetic resonance）6.1.150核磁共振（nuclear magnetic resonance）6.1.151霍尔效应（hall effect）6.1.152 波尔磁子（Bohr magneton）6.1.153 质子旋磁比（Proton gyro magnetic ratio）6.1.154 磁通量子（F1ux quantum (F1uxon)）6.2 电学计量6.2.1.1直流电压基准(Primary Standard of DC V oltage)6.2.1.2直流电动势基准(Primary Standard of DC Electromotive Force)6.2.1.3直流电阻基准(Primary Standard of DC Resistance)6.2.1.4电容基准(Primary Standard of Capacitance)6.2.1.5电容器损耗因数基准(Primary Standard of Dissipation Factor)6.2.1.6电感基准(Primary Standard of Inductance)6.2.1.7交流电流基准(Primary Standard of AC Current)6.2.1.8交流电压基准(Primary Standard of AC V oltage)6.2.1.9交流功率基准(Primary Standard of AC Power)6.2.1.10工频电能基准(Primary Standard of AC Energy at Industrial Frequency)6.2.1.11磁感应强度基准(Primary Standard of Magnetic Flux Density)6.2.1.12数字阻抗电桥标准(Standard for LCR meter)6.2.1.13数字多用表检定装置(Standard of Multimeter)6.2.1.14超导强磁场标准(Standard of Supper Conducting High Magnetic Field)6.2.1.15非铁磁金属电导率标准(Standard of Conductivity for Nonferrous Metals)6.2.1.16模/数、数/模转换测量标准（Standard of ADC and DAC）6.2.1.17标准电池(standard cell)6.2.1.18固态电压标准(solid state voltage standard)6.2.1.19标准电阻(standard resistor)6.2.1.20计算电容(cross capacitor)6.2.1.21感应分压器(inductive voltage divider)6.2.1.22分流器(shunt)6.2.1.23直流电流比较仪(direct current comparator)6.2.1.25多功能校准源(multifunction calibrator)6.2.1.26数字阻抗电桥(LCR meter)6.2.1.27 电压表（voltmeter）6.2.1.28 电流表(amperometer)6.2.1.29 电阻表(ohnneter)6.2.1.30 功率表(Watt meter)6.2.1.31 电能表(kWh meter)6.2.2电学计量常用测量方法6.2.2.1 直接测量（法）(direct (method of) measurement)6.2.2.3组合测量（法）(combination (method of) measurement)6.2.2.4 比较测量（法）(comparison (method of) measurement)6.2.2.5 零值测量（法）(null (method of) measurement)6.2.2.6 差值测量（法）(differential (method of) measurement)6.2.2.7 替代测量（法）(substitution (method of) measurement)6.2.2.8 不完全替代法(semi-substitution method of measurement)6.2.2.9 内插测量（法）(interpolation (method of) measurement)6.2.2.10 互补测量（法）(complementary (method of) measurement)6.2.2.11 差拍测量（法）(beat (method of) measurement)6.2.2.12 谐振测量（法）(resonance (method of) measurement)6.2.2.13 模数转换(analogue to digital conversion)6.2.2.14 数模转换(digital to analogue conversion)6.2.2.15 静电屏蔽(electrostatic screen)6.2.2.16 磁屏蔽(magnetic screen)6.2.2.17 泄漏电流(leakage current)6.2.2.18 电位屏蔽(potential screen)6.2.2.19 等电位屏蔽(equip—potential screen)6.2.2.20 无定向结构(astatic construction)6.2.2.21交流-直流转换(AC-DC conversion) 6.2.2.22交流-直流转换器<AC-DC converter) 6.2.2.23交流-直流比较仪(AC-DC comparator)6.2.2.24热电变换器(thermal converter)6.2.2.25 共模电压(common mode voltage)6.2.2.26 串模电压(series mode voltage)6.2.2.27 共模抑制比(common mode rejection ratio ———CMRR)6.2.2.28 串模抑制比(series mode rejection ratio——SMRR)6.2.2.29 非对称输入(asymmetrical input) 6.2.2.30 非对称输出(asymmetrical output) 6.2.2.31 对称输入(symmetrical input)6.2.2.32 对称输出(symmetrical output)6.2.2.33 差分输入电路(differential input circuit)6.2.2.34 接地输入电路(earthed input circuit 或grounded input)6.2.2.35 接地输出电路(earthed output circuit或grounded output)6.2.2.36 浮置输入电路(floating input circuit)6.2.2.37 浮置输出电路(floating output circuit)6.2.3.1 模拟（测量）仪表（analogue (measuring) instrument）模拟指示仪表（analogue indicating instrument）6.2.3.2 数字（测量）仪表（digital (measuring) instrument）6.2.3.3 热电系仪表（electrothermal instrument）6.2.3.4 双金属系仪表（bimetallic instrument）6.2.3.5 热偶式仪表（thermocouple instrument）6.2.3.6 整流式仪表（rectifier instrument）6.2.3.7 振簧系仪表（vibrating reed instrument）6.2.3.8 多用表、万用表（multimeter）6.2.3.9（测量）电桥（(measuring) bridge）6.2.3.10（测量）电位差计（(measuring) potentiometer）6.2.3.11 分压器（voltage divider）6.2.3.12 比较仪（comparator）6.2.3.13 指针式仪表（pointer instrument）6.2.3.14 光标式仪表（instrument with optical index）6.2.3.15 动标度仪表（moving-scale instrument）6.2.3.16 影条式仪表（shadow column instrument）6.2.3.17 静电系仪表（electrostatic instrument）6.2.3.18 磁电系仪表（(permanent magnet) moving-coil instrument）6.2.3.19 动磁系仪表（moving magnet instrument）6.2.3.20 电磁系仪表（moving-iron instrument）6.2.3.21 电动系仪表（electrodynamic instrument）6.2.3.22 铁磁电动系仪表（ferrodynamic instrument）6.2.3.23 感应系仪表（induction instrument）。

Common M ode F ilter D esign G uideIntroductionThe selection of component values for common mode filters need not be a difficult and confusing process. The use of standard filter alignments can be utilized to achieve a relatively simple and straightforward design process, though such alignments may readily be modified to utilize pre-defined component values.GeneralLine filters prevent excessive noise from being conducted between electronic equipment and the AC line; generally, the emphasis is on protecting the AC line. Figure 1 shows the use of a common mode filter between the AC line (via impedance matching circuitry) and a (noisy) power con-verter. The direction of common mode noise (noise on both lines occurring simultaneously referred to earth ground) is from the load and into the filter, where the noise common to both lines becomes sufficiently attenuated. The result-ing common mode output of the filter onto the AC line (via impedance matching circuitry) is then negligible.Figure 1.Generalized line filteringThe design of a common mode filter is essentially the design of two identical differential filters, one for each of the two polarity lines with the inductors of each side coupled by a single core:L2Figure 2.The common mode inductorFor a differential input current ( (A) to (B) through L1 and (B) to (A) through L2), the net magnetic flux which is coupled between the two inductors is zero.Any inductance encountered by the differential signal is then the result of imperfect coupling of the two chokes; they perform as independent components with their leak-age inductances responding to the differential signal: the leakage inductances attenuate the differential signal. When the inductors, L1 and L2, encounter an identical signal of the same polarity referred to ground (common mode signal), they each contribute a net, non-zero flux in the shared core; the inductors thus perform as indepen-dent components with their mutual inductance respond-ing to the common signal: the mutual inductance then attenuates this common signal.The First Order FilterThe simplest and least expensive filter to design is a first order filter; this type of filter uses a single reactive component to store certain bands of a spectral energy without passing this energy to the load. In the case of a low pass common mode filter, a common mode choke is the reactive element employed.The value of inductance required of the choke is simply the load in Ohms divided by the radian frequency at and above which the signal is to be attenuated. For example, attenu-ation at and above 4000 Hz into a 50⏲ load would require a 1.99 mH (50/(2π x 4000)) inductor. The resulting common mode filter configuration would be as follows:50Ω1.99 mHFigure 3.A first order (single pole) common mode filter The attenuation at 4000 Hz would be 3 dB, increasing at 6 dB per octave. Because of the predominant inductor dependence of a first order filter, the variations of actual choke inductance must be considered. For example, a ±20% variation of rated inductance means that the nominal 3 dB frequency of 4000 Hz could actually be anywhere in the range from 3332 Hz to 4999 Hz. It is typical for the inductance value of a common mode choketo be specified as a minimum requirement, thus insuring that the crossover frequency not be shifted too high.However, some care should be observed in choosing a choke for a first order low pass filter because a much higher than typical or minimum value of inductance may limit the choke’s useful band of attenuation.Second Order FiltersA second order filter uses two reactive components and has two advantages over the first order filter: 1) ideally, a second order filter provides 12 dB per octave attenuation (four times that of a first order filter) after the cutoff point,and 2) it provides greater attenuation at frequencies above inductor self-resonance (See Figure 4).One of the critical factors involved in the operation of higher order filters is the attenuating character at the corner frequency. Assuming tight coupling of the filter components and reasonable coupling of the choke itself (conditions we would expect to achieve), the gain near the cutoff point may be very large (several dB); moreover, the time response would be slow and oscillatory. On the other hand, the gain at the crossover point may also be less than the presumed -3 dB (3 dB attenuation), providing a good transient response, but frequency response near and below the corner frequency could be less than optimally flat.In the design of a second order filter, the damping factor (usually signified by the Greek letter zeta (ζ )) describes both the gain at the corner frequency and the time response of the filter. Figure (5) shows normalized plots of the gain versus frequency for various values of zeta.Figure 4.Analysis of a second order (two pole) common modelow pass filterThe design of a second order filter requires more care and analysis than a first order filter to obtain a suitable response near the cutoff point, but there is less concern needed at higher frequencies as previously mentioned.A ≡ ζ = 0.1;B ≡ ζ = 0.5;C ≡ ζ = 0.707;D ≡ ζ = 1.0;E ≡ ζ = 4.0Figure 5.Second order frequency response for variousdamping f actors (ζ)As the damping factor becomes smaller, the gain at the corner frequency becomes larger; the ideal limit for zero damping would be infinite gain. The inherent parasitics of real components reduce the gain expected from ideal components, but tailoring the frequency response within the few octaves of critical cutoff point is still effectively a function of ideal filter parameters (i.e., frequency, capaci-tance, inductance, resistance).L0.1W n1W n 10W nRadian Frequency,WG a i n (d B )V s V s LR s LCs LC j L R j LC LR LCCMout CMin L L n n n L ()()=++=−+⎛⎝⎜⎞⎠⎟=+−⎛⎝⎜⎞⎠⎟≡≡≡≡111111212222ωωζωωωωωωζradian frequencyR the noise load resistance LFor some types of filters, the design and damping char-acteristics may need to be maintained to meet specific performance requirements. For many actual line filters,however, a damping factor of approximately 1 or greater and a cutoff frequency within about an octave of the calculated ideal should provide suitable filtering.The following is an example of a second order low pass filter design:1)Identify the required cutoff frequency:For this example, suppose we have a switching power supply (for use in equipment covered by UL478) that is actually 24 dB noisier at 60 KH z than permissible for the intended application. For a second order filter (12dB/octave roll off) the desired corner frequency would be 15 KHz.2)Identify the load resistance at the cutoff frequency:Assume R L = 50 Ω3)Choose the desired damping factor:Choose a minimum of 0.707 which will provide 3 dB attenuation at the corner frequency while providing favorable control over filter ringing.4)Calculate required component values:Note:Damping factors much greater than 1 may causeunacceptably high attenuation of lower frequen-cies whereas a damping factor much less than 0.707 may cause undesired ringing and the filter may itself produce noise.Third Order FiltersA third order filter ideally yields an attenuation of 18 dB per octave above the cutoff point (or cutoff points if the three corner frequencies are not simultaneous); this is the prominently positive aspect of this higher order filter. The primary disadvantage is cost since three reactive compo-nents are now required. H igher than third order filters are generally cost-prohibitive.Figure 6.Analysis of a third order (three pole) low pass filter where ω1, ω2 and ω4 occur at the same -3dB frequency of ω05)Choose available components:C = 0.05 µF (Largest standard capacitor value that will meet leakage current requirements for UL478/CSA C22.2 No. 1: a 300% decrease from design)L = 2.1 mH (Approx. 300% larger than design to compensate for reduction or capacitance: Coilcraft standard part #E3493-A)6)Calculate actual frequency, damping factor, and at-tenuation for components chosen:ζ = 2.05 (a damping factor of about 1 or more is acceptible)Attenuation = (12 dB/octave) x 2 octaves = 24 dB 7)The resulting filter is that of figure (4) with:L = 2.1 mH; C = 0.05 µF; R L = 50 ΩL 1L 2VCMout s VCMin s R R L s R L s sC R L s sC R L s L L s L s sC L L R s L Cs L L C R s L L L L L L L()()()()=+⎛⎝⎜⎞⎠⎟+++++⎛⎝⎜⎜⎜⎜⎞⎠⎟⎟⎟⎟=++++222121*********11Butterworth →+++112212233s s s n n n ωωω()()L L R R L L L n n L 12111222+==+ωω;()L L C n 1n2C =2;ωω2211414=.L L L L n n n 12L n3n2L2n2L2C R =1;R R ωωωωωω33224422===ωπωζωμn n n Lf C L L R L =====294248070727502rad /sec =1Hn .1215532πLC=Hz (very nearly 15KHz)The design of a generic filter is readily accomplished by using standard alignments such as the Butterworth (“maxi-mally flat”) alignments. Figure (6) shows the general analysis and component relationships to the Butterworth alignments for a third order low pass filter. Butterworth alignments provide an inherent ζ of 0.707 and a -3 dB point at the crossover frequency. The Butterworth alignments for the first three orders of low pass filters are shown in Figure (7).The design of a line filter need not obey the Butterworth alignments precisely (although such alignments do pro-vide a good basis for design); moreover, because of leakage current limits placed upon electronic equipment (thus limiting the amount of filter capacitance to ground),adjustments to the alignments are usually required, but they can be executed very simply as follows:1)First design a second order low pass with ζ ≥ 0.52)Add a third pole (which has the desired corner fre-quency) by cascading a second inductor between the second order filter and the noise load:L = R/ (2 π f c )Where f c is the desired corner frequency.Design ProcedureThe following example determines the required compo-nent values for a third order filter (for the same require-ments as the previous second order design example).1)List the desired crossover frequency, load resistance:Choose f c = 15000 Hz Choose R L = 50 Ω2)Design a second order filter with ζ = 0.5 (see second order example above):3)Design the third pole:R L /(2πf c ) = L 250/(2π15000) = 0.531 mH4)Choose available components and check the resulting cutoff frequency and attenuation:L2 = 0.508 mH (Coilcraft #E3506-A)f n= R/(2πL 1 )= 15665 HzAttenuation at 60 KHZ: 24 dB (second order filter) +2.9 octave × 6 = 41.4 dB5)The resulting filter configuration is that of figure (6)with:L 1 = 2.1 mH L 2 = 0.508 mH R L = 50 ΩConclusionsSpecific filter alignments may be calculated by manipu-lating the transfer function coefficients (component val-ues) of a filter to achieve a specific damping factor.A step-by-step design procedure may utilize standard filter alignments, eliminating the need to calculate the damping factor directly for critical filtering. Line filters,with their unique requirements, yet non-critical character-istics, are easily designed using a minimum allowable damping factor.Standard filter alignments assume ideal filter compo-nents; this does not necessarily hold true, especially at higher frequencies. For a discussion of the non-ideal character of common mode filter inductors refer to the application note “Common Mode Filter Inductor Analysis,”available from Coilcraft.Figure 7.The first three order low pass filters and their Butterworth alignmentse i +–e O +–R LL 2Ce i +–e O +–R LL 1Ce i +–e O +–R LL 1L 2Filter SchematicFilter Transfer FunctionButterworthAlignmentFirst OrderSecond OrderThird Ordere e Ls R o iL =+11e e LCs Ls R oi L=++112e e L L R s L Cs L L s R o iLL =++++111231212()e e s o in=+11ωe e LCs Ls R oiL =++112e e s s so i n n n =+++122133221ωωω。

前言Tune Matching的方法有许多，有利用单独供电给PA，直接在Active情况下Tune Matching的方式[1]，但是这方法要有两个条件:1.能够正常通话2.能进入非信令模式然而Tune Matching的工作，多半都是在第一版PCB就要完成(因为第二版PCB 就要直接送认证)，但是依个人经验，通常第一版PCB，软件可能尚未Ready，正常通话? 进入非信令模式? 再等等呗。

因此个人较偏好利用Passive方式Tune Matching，你只要有板子就能进行，不必等到软件Ready。

由于GSM跟WCDMA是手机的核心，故个人以这两个功能的Tx/Rx Matching来做说明。

基本原理:最理想情况，当然是希望Source端的输出阻抗为50奥姆，传输线的阻抗为50奥姆，Load端的输入阻抗也是50奥姆，一路50奥姆下去，这是最理想的。

但是，板厂的制程，在Trace的线宽，以及对地间距，一定会有误差，这导致Trace的阻抗，未必是50奥姆，所以要靠Matching把阻抗Tune到50奥姆。

所以通常就算对于阻抗控制再有信心，也会留Dummy pad，以备不时之需。

Matching步骤:先把落地组件拔掉，串联组件用0奥姆电阻，目的是要知道PCB Trace最原始的阻抗为多少，接下来才能利用Smith Chart跟Matching组件，把阻抗Tune到50奥姆。

这样比较省事。

Q. 我可以直接用焊锡Short，来代替0奥姆电阻吗? 这样比较省事答案是不行，因为虽然以电路观点，都是Short，但是以高频观点，利用焊锡这种Distributed方式，会有寄生效应，连带使得你量出来的阻抗会不准。

零件换好后，先把网络分析仪做校正，再将铜管作Port extension，如此便可开始量阻抗。

我们发现PCB Trace最原始的负载阻抗为(40.6-13j)奥姆，接下来就是利用Smith Chart，将负载阻抗Tune到50奥姆。

Understanding the Fundamental Principles of Vector Network AnalysisTable of ContentsIntroduction (3)Measurements in Communications Systems (3)Importance of Vector Measurements (5)The Basis of Incident and Reflected Power (6)The Smith Chart (6)Power Transfer Conditions (7)Network Analysis Terminology (10)Measuring Group Delay (12)Network Characterization (13)Related Literature (15)IntroductionNetwork analysis is the process by which designers and manufacturers measure the electrical performance of the components and circuits used in more complex systems. When these systems are conveying signals with information content, we are most concerned with getting the signal from one point to another with maximum efficiency and minimum distortion. Vector network analysis is a method of accurately characterizing such components by measuring their effect on the amplitude and phase of swept-frequency and swept-power test signals. In this application note, the fundamental principles of vector network analysiswill be reviewed. The discussion includes the common parameters that can be measured, including the concept of scattering parameters (S-parameters). RF fun-damentals such as transmission lines and the Smith chart will also be reviewed. Agilent Technologies offers a wide range of portable and benchtop vector network analyzers for characterizing components from DC to 110 GHz. These instruments are available with a wide range of options to simplify testing in the field, laboratory, and production environments.Measurements in Communications SystemsIn any communications system, the effect of signal distortion must be consid-ered. While we generally think of the distortion caused by nonlinear effects (for example, when intermodulation products are produced from desired carrier signals), purely linear systems can also introduce signal distortion. Linear systems can change the time waveform of signals passing through them by altering the amplitude or phase relationships of the spectral components that make up the signal.Let’s examine the difference between linear and nonlinear behavior more closely. Linear devices impose magnitude and phase changes on input signals (Figure 1). Any sinusoid appearing at the input will also appear at the output, and at the same frequency. No new signals are created. Both active and passive nonlinear devices can shift an input signal in frequency or add other frequency components, such as harmonic and spurious signals. Large input signals can drive normally linear devices into compression or saturation, causing nonlinear operation.Figure 1. Linear versus nonlinear behaviorFor linear distortion-free transmission, the amplitude response of the device under test (DUT) must be flat and the phase response must be linear over the desired bandwidth. As an example, consider a square-wave signal rich in high-frequency components passing through a bandpass filter that passes selected frequencies with little attenuation while attenuating frequencies outside of the passband by varying amounts.Even if the filter has linear phase performance, the out-of-band componentsof the square wave will be attenuated, leaving an output signal that, in this example, is more sinusoidal in nature (Figure 2).If the same square-wave input signal is passed through a filter that only inverts the phase of the third harmonic, but leaves the harmonic amplitudes the same, the output will be more impulse-like in nature (Figure 3). While this is true for the example filter, in general, the output waveform will appear with arbitrary distortion, depending on the amplitude and phase nonlinearities.Figure 2. Magnitude variation with frequencyFigure 3. Phase variation with frequencyFigure 4. Nonlinear induced distortionNonlinear devices also introduce distortion (Figure 4). For example, if an ampli-fier is overdriven, the output signal clips because the amplifier is saturated. The output signal is no longer a pure sinusoid, and harmonics are present at multiples of the input frequency. Passive devices may also exhibit nonlinear behavior at high power levels, a good example of which is an L-C filter that uses inductors with magnetic cores. Magnetic materials often exhibit hysteresis effects that are highly nonlinear.Efficient transfer of power is another fundamental concern in communications systems. In order to efficiently convey, transmit or receive RF power, devices such as transmissions lines, antennas and amplifiers must present the proper impedance match to the signal source. Impedance mismatches occur when the real and imaginary parts of input and output impedances are not ideal between two connecting devices.Importance of Vector MeasurementsMeasuring both magnitude and phase of components is important for several reasons. First, both measurements are required to fully characterize a linear network and ensure distortion-free transmission. To design efficient matching networks, complex impedance must be measured. Engineers developing models for computer-aided-engineering (CAE) circuit simulation programs require magnitude and phase data for accurate models.In addition, time-domain characterization requires magnitude and phase information in order to perform an inverse-fourier transform. Vector error correction, which improves measurement accuracy by removing the effectsof inherent measurement-system errors, requires both magnitude and phase data to build an effective error model. Phase-measurement capability is very important even for scalar measurements such as return loss, in order to achieve a high level of accuracy (see Applying Error Correction to Network Analyzer Measurements, Agilent application note 1287-3).The Basis of Incident and Refl ected PowerIn its fundamental form, network analysis involves the measurement of incident, reflected, and transmitted waves that travel along transmission lines. Using optical wavelengths as an analogy, when light strikes a clear lens (the incident energy), some of the light is reflected from the lens surface, but most of it continues through the lens (the transmitted energy) (Figure 5). If the lens has mirrored surfaces, most of the light will be reflected and little or none will pass through it.While the wavelengths are different for RF and microwave signals, the principle is the same. Network analyzers accurately measure the incident, reflected, and transmitted energy, e.g., the energy that is launched onto a transmission line, reflected back down the transmission line toward the source (due to impedence mismatch), and successfully transmitted to the terminating device (such as an antenna).Figure 5. Lightwave analogy to high-frequency device characterizationThe Smith ChartThe amount of reflection that occurs when characterizing a device depends on the impedance that the incident signal “sees.” Since any impedance can be represented with real and imaginary parts (R + jX or G + jB), they can be plotted on a rectilinear grid known as the complex impedance plane. Unfortunately, an open circuit (a common RF impedence) appears at infinity on the real axis, and therefore cannot be shown.The polar plot is useful because the entire impedance plane is covered. However, instead of plotting impedance directly, the complex reflection coef-ficient is displayed in vector form. The magnitude of the vector is the distance from the center of the display, and phase is displayed as the angle of vector referenced to a flat line from the center to the right-most edge. The drawback of polar plots is that impedance values cannot be read directly from the display.Since there is a one-to-one correspondence between complex impedance and reflection coefficient, the positive real half of the complex impedance plane can be mapped onto the polar display. The result is the Smith chart. All values of reactance and all positive values of resistance from 0 to infinity fall within the outer circle of the Smith chart (Figure 6).On the Smith chart, loci of constant resistance appear as circles, while loci of constant reactance appear as arcs. Impedances on the Smith chart are always normalized to the characteristic impedance of the component or system of inter-est, usually 50 ohms for RF and microwave systems and 75 ohms for broadcast and cable-television systems. A perfect termination appears in the center of the Smith chart.Power Transfer ConditionsA perfectly matched condition must exist at a connection between two devices for maximum power transfer into a load, given a source resistance of R S and a load resistance of R L . This condition occurs when R L = R S , and is true whether the stimulus is a DC voltage source or a source of RF sine waves (Figure 7). When the source impedance is not purely resistive, maximum power transfer occurs when the load impedance is equal to the complex conjugate of the source impedance. This condition is met by reversing the sign of the imaginary part of the impedance. For example, if R S = 0.6 + j 0.3, then the complex conjugate is R S * = 0.6 – j 0.3.The need for efficient power transfer is one of the main reasons for the use of transmission lines at higher frequencies. At very low frequencies (with much larger wavelengths), a simple wire is adequate for conducting power. The resis-tance of the wire is relatively low and has little effect on low-frequency signals. The voltage and current are the same no matter where a measurement is made on the wire.At higher frequencies, wavelengths are comparable to or smaller than the length of the conductors in a high-frequency circuit, and power transmission can be thought of in terms of traveling waves. When the transmission line is terminated in its characteristic impedance, maximum power is transferred to the load. When the termination is not equal to the characteristic impedance, that part of the signal that is not absorbed by the load is reflected back tothe source.If a transmission line is terminated in its characteristic impedance, no reflected signal occurs since all of the transmitted power is absorbed by the load (Figure 8). Looking at the envelope of the RF signal versus distance along the transmission line shows no standing waves because without reflections, energy flows in only one direction.Figure 7. Power transferFigure 8. Transmission line terminated with ZWhen the transmission line is terminated in a short circuit (which can sustain no voltage and therefore dissipates zero power), a reflected wave is launched back along the line toward the source (Figure 9). The reflected voltage wave must be equal in magnitude to the incident voltage wave and be 180 degrees out of phase with it at the plane of the load. The reflected and incident waves are equal in magnitude but traveling in the opposite directions.If the transmission line is terminated in an open-circuit condition (which can sustain no current), the reflected current wave will be 180 degrees out of phase with the incident current wave, while the reflected voltage wave will be in phase with the incident voltage wave at the plane of the load. This guarantees that the current at the open will be zero. The reflected and incident current waves are equal in magnitude, but traveling in the opposite directions. For both the short and open cases, a standing wave pattern is set up on the transmission line. The voltage valleys will be zero and the voltage peaks will be twice the incident voltage level.If the transmission line is terminated with say a 25-ohm resistor, resulting ina condition between full absorption and full reflection, part of the incident power is absorbed and part is reflected. The amplitude of the reflected voltage wave will be one-third that of the incident wave, and the two waves will be 180 degrees out of phase at the plane of the load. The valleys of the standing-wave pattern will no longer be zero, and the peaks will be less than those of the short and open cases. The ratio of the peaks to valleys will be 2:1.The traditional way of determining RF impedance was to measure VSWR using an RF probe/detector, a length of slotted transmission line, and a VSWR meter. As the probe was moved along the transmission line, the relative position and values of the peaks and valleys were noted on the meter. From these measurements, impedance could be derived. The procedure was repeated at different frequencies. Modern network analyzers measure the incident and reflected waves directly during a frequency sweep, and impedance results can be displayed in any number of formats (including VSWR).Figure 9. Transmission line terminated with short, openNetwork Analysis TerminologyNow that we understand the fundamentals of electromagnetic waves, we must learn the common terms used for measuring them. Network analyzer terminology generally denotes measurements of the incident wave with the R or reference channel. The reflected wave is measured with the A channel, and the transmitted wave is measured with the B channel (Figure 10). With the amplitude and phase information in these waves, it is possible to quantify the reflection and transmission characteristics of a DUT. The reflection and transmis-sion characteristics can be expressed as vector (magnitude and phase), scalar (magnitude only), or phase-only quantities. For example, return loss is a scalar measurement of reflection, while impedance is a vector reflection measurement. Ratioed measurements allow us to make reflection and transmission measure-ments that are independent of both absolute power and variations in source power versus frequency. Ratioed reflection is often shown as A/R and ratioed transmission as B/R, relating to the measurement channels in the instrument. Figure 10. Common terms for high-frequency device characterizationThe most general term for ratioed reflection is the complex reflection coef-ficient, G or gamma (Figure 11). The magnitude portion of G is called r or rho. The reflection coefficient is the ratio of the reflected signal voltage level to the incident signal voltage level. For example, a transmission line terminated inits characteristic impedance Zo ,will have all energy transferred to the load soV refl = 0 and r = 0. When the impedance of the load, ZLis not equal to the char-acteristic impedance, energy is reflected and r is greater than zero. When the load impedance is equal to a short or open circuit, all energy is reflected and r = 1. As a result, the range of possible values for r is 0 to 1.Figure 11. Refl ection parametersReturn loss is a way to express the reflection coefficient in logarithmic terms (decibels). Return loss is the number of decibels that the reflected signal is below the incident signal. Return loss is always expressed as a positive number and varies between infinity for a load at the characteristic impedance and 0 dB for an open or short circuit. Another common term used to express reflection is voltage standing wave ratio (VSWR), which is defined as the maximum value of the RF envelope over the minimum value of the RF envelope. It is related to r as (1 + r)/(1 – r). VSWR ranges from 1 (no reflection) to infinity (full reflection). The transmission coefficient is defined as the transmitted voltage divided by the incident voltage (Figure 12). If the absolute value of the transmitted voltage is greater than the absolute value of the incident voltage, a DUT or system is said to have gain. If the absolute value of the transmitted voltage is less than the absolute value of the incident voltage, the DUT or system is said to have attenuation or insertion loss. The phase portion of the transmission coefficient is called insertion phase.Figure 12. Transmission parametersDirect examination of insertion phase usually does not provide useful information. This is because the insertion phase has a large (negative) slope with respect to frequency due to the electrical length of the DUT. The slope is proportional to the length of the DUT. Since it is only deviation from linear phase that causes distor-tion in communications systems, it is desirable to remove the linear portion of the phase response to analyze the remaining nonlinear portion. This can be done by using the electrical delay feature of a network analyzer to mathematically cancel the average electrical length of the DUT. The result is a high-resolution display of phase distortion or deviation from linear phase (Figure 13).Figure 13. Deviation from linear phaseMeasuring Group DelayAnother useful measure of phase distortion is group delay (Figure 14). This parameter is a measure of the transit time of a signal through a DUT versus frequency. Group delay can be calculated by differentiating the DUT’s phase response versus frequency. It reduces the linear portion of the phase response to a constant value, and transforms the deviations from linear phase into deviations from constant group delay, (which causes phase distortion in communications systems). The average delay represents the average signal transit time through a DUT.Figure 14. What is group delay?Depending on the device, both deviation from linear phase and group delay may be measured, since both can be important. Specifying a maximum peak-to-peak phase ripple in a device may not be sufficient to completely characterize it, since the slope of the phase ripple depends on the number of ripples that occur per unit of frequency. Group delay takes this into account because it is the differen-tiated phase response. Group delay is often a more easily interpreted indication of phase distortion (Figure 15).Figure 15. Why measure group delay?Network CharacterizationIn order to completely characterize an unknown linear two-port device, we must make measurements under various conditions and compute a set of parameters. These parameters can be used to completely describe the electrical behavior of our device (or network), even under source and load conditions other than when we made our measurements. Low-frequency device or network characterization is usually based on measurement of H, Y, and Z parameters. To do this, the total voltage and current at the input or output ports of a device or nodes of a network must be measured. Furthermore, measurements must be made with open-circuit and short-circuit conditions.Since it is difficult to measure total current or voltage at higher frequencies,S-parameters are generally measured instead (Figure 16). These parameters relate to familiar measurements such as gain, loss, and reflection coefficient. They are relatively simple to measure, and do not require connection of undesirable loads to the DUT. The measured S-parameters of multiple devices can be cascaded to predict overall system performance. S-parameters are read-ily used in both linear and nonlinear CAE circuit simulation tools, and H, Y, and Z parameters can be derived from S-parameters when necessary.The number of S-parameters for a given device is equal to the square of the number of ports. For example, a two-port device has four S-parameters. The numbering convention for S-parameters is that the first number following theS is the port at which energy emerges, and the second number is the port atwhich energy enters. So S21 is a measure of power emerging from Port 2 as aresult of applying an RF stimulus to Port 1. When the numbers are the same(e.g. S11), a reflection measurement is indicated.Forward S-parameters are determined by measuring the magnitude and phase of the incident, reflected, and transmitted signals when the output is terminated in a load that is precisely equal to the characteristic impedance of the test system. In the case of a simple two-port network, S 11 is equivalent to the input complex reflection coefficient or impedance of the DUT, while S 21 is the forward complex transmission coefficient. By placing the source at the output port of the DUT and terminating the input port in a perfect load, it is possible to measure the other two (reverse) S-parameters. Parameter S 22 is equivalent to the output complex reflection coefficient or output impedance of the DUT while S 12 is the reverse complex transmission coefficient (Figure 17).Figure 16. Limitations of H, Y, and Z parameters (Why use S-parameters?)Figure 17. Measuring S-parametersFor more information on AgilentTechnologies’ products, applications or services, please contact your local Agilent office. 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To keep you competitive, we continually invest in tools andprocesses that speed up calibration and repair and reduce your cost of ownership. You can also use Infoline Web Services to manage equipment and services more effectively. By sharing our measurement and service expertise, we help you create the products that change our world./find/advantageservicesAgilent Email Updates/find/emailupdatesGet the latest information on the products and applications you select./qualityQuality Management SystemQuality Management Sys ISO 9001:2008DEKRA Certified Related LiteratureExploring the Architectures of Network Analyzers , Application Note 1287-2, Literature number 5965-7708E Applying Error Correction toNetwork Analyzer Measurements , Application Note 1287-3, Literature number 5965-7709E Network Analyzer Measurements: Filter and Amplifier Examples, Application Note 1287-4,Literature number 5965-7710E。