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小型宽波束圆极化天线及馈电网络丁轲佳;吕善伟;张岩【摘要】提出一种新型的小型宽波束圆极化天线及馈电网络。
天线由四个独立辐射单元顺次旋转90。
组合而成，辐射单元采用平面印刷电路工艺加工、使用容性加载及短路点调配方式实现小型化；使用变形的分支线定向耦合器与移相电路组合，实现具有功率分配与正交移相功能的馈电网络。
经理论分析、软件仿真及实测表明：天线高度小于0．12λ，3dB波束宽度大于120°，6dB轴比波束宽度大于180°，辐射特性在15％相对带宽内稳定；天线的驻波比（VSWR）小于2．0的相对带宽为16％。
该小型宽波束圆极化天线采用印刷电路工艺制作，结构紧凑，加工成本低、精度高，易于与微波电路集成，其结构也适用于S波段以上更高频段。
%A kind of novel compact wide-beam circular polarization antenna and its feed network are presented. The antenna is composed of four independent radiation units which rotate 90° about the center separately. The miniaturization of the anten- na is realized by distributed capacitive load and short point. The feed network is composed of directional coupler and microstrip lines to achieve power division and quartered phase shift. The analysis, simulation and measured results indicate that the antenna＇s height is less than 0.12λ, the antenna＇s beam width is greater than 120°,the beam width of axial ratio less than 6dB is greater than 180°,and the char- acters mentioned above can be obtained in the frequency range over 15° band wi dth,the relative band width of VSWR〈2.0 is 16%. The antenna is fabricated by PCB technics and could be integrated to microwave circuit conveniently. The fabri- cation has low cost and highprecision. The structure could also be applied to fre- quency range higher than S band and has broad prospects of application.【期刊名称】《电波科学学报》【年(卷),期】2012(027)004【总页数】6页(P680-684,746)【关键词】宽波束;圆极化;小型化;轴比【作者】丁轲佳;吕善伟;张岩【作者单位】北京航空航天大学,北京100191;北京航空航天大学,北京100191;北京航空航天大学,北京100191【正文语种】中文【中图分类】TN011.4引言伴随卫星通信快速发展，卫星定位与导航、卫星广播通信等技术在军民用领域获得广泛应用。
图1 梯形多缝天线示意图1 梯形多缝天线简介梯形多缝天线是一种具有较好的宽频带工作性能的多缝天线，其结构如图1所示。
梯形多缝天线利用多条辐射缝隙工作频段叠加实现宽频工作，它由3条以上直线缝隙组成，从上到下，直线缝隙的长度逐渐增加。
每条直线缝隙的长度不同，工作频带不同，多条直线缝隙的辐射叠加，可以形成一个工作带宽较大的工作频带。
2 六边形光子晶体结构简介六边形光子晶体结构如图2所示。
把一个金属六边形结构分为12个直角三角形，在每个直角三角形的中心形阵列结构设计阵列天线可以保证天线具有超宽频带工作特性。
4 梯形多缝-六边形阵列复合超宽频带天线结构设计在设计中，使用低损耗微波陶瓷基板作为天线的介质基板，其相对介电常数为50，基板的形状为矩形寸是40 mm×41.6 mm，厚度为1 mm的正面贴覆有天线的梯形多缝-六边形阵列复合辐射贴片，其结构如图4所示。
微波陶瓷基板的背面贴覆有天线的光子晶体-六边形阵列复合接地板图2 六边形光子晶体示意图图3 六边形阵列结构的排列方式示意图图4 梯形多缝-六边形阵列复合辐射贴片结构示意图的长度为金属直角三角形各边的长度的一半。
梯形多缝-六边形阵列复合天线很好地将梯形多缝天线和六边形阵列结构的优点结合起来，梯形多缝天线通过多条不同长度的缝隙的辐射叠加，保证天线有较高的辐射强度和较大的工作带宽。
六边形阵列结构的完美对称，使射频电流在天线内部均匀分布，增大了天线的工作带宽并保证天线在各个工作频段的辐射性能均匀稳定。
光子晶体-六边形阵列复合接地板利用产生的光子带隙进一步拓展了天线的工作频段，使天线具有优异的超宽频段工作性能。
5 梯形多缝-六边形阵列复合超宽频带天线4.400 GHz～4.500 GHz、4.800 GHz～4.990 GHz、5.725 GHz～5.875 GHz、3.100 GHz～10.600 GHz、11.700 GHz～12.200 GHz等第二代至第五代移动通信所有制式所有工作频段、射频识别系统工作频段、超宽带系统工作频段、移动数字电视系统工作频段。
通信技术终端的宽带扁平状双频圆极化微带天线，李晓鹏1，李成钢2，蔡惠萍广州中海达卫星导航技术股份有限公司，广东广州511400；2.广州市中海达测绘仪器有限公司，广东终端的宽带扁平状双频圆极化微带天线。
通过在天线辐射单元基板外围设置短路加载振子，不仅可以降低天线谐振频率实现天线的小型化设计，而且可有效提升天线辐射增益和工作带宽。
双频天线单元分别采用四馈点馈电方式，使得天线拥有稳定且可靠的相位中心偏差值和良好的圆极化特性。
针对该天线设计模型，使用仿真软件对天线进行仿真，可以得出天线在高频段（1.1661.607 GHz）内的各项指标表现较优。
通过制作实物样机进行实际测量，结果表明该天线在上述高低频双1.9 dBi，对应高低频段内中心频点的辐射增益大于180°。
因此，该天线能够较好满足目前GNSS短路加载；小型化；宽频段；相位中心；高增益Wide-Band Flat Dual-Band Circularly Polarized Microstrip Antenna for GNSSApplications，LI Chenggang2，CAI Huiping.Hi-Target Navigation Tech Co.，Ltd.，Guangzhou（1）（2）r为相对天线馈电点的位置可由公式粗略估算，然后通。
给出输入电阻与馈点（3）为天线馈点距离中心点位置；h0为平均电场强度；P为辐图1 天线结构俯视图2 仿真与实测分析本文采用专业电磁仿真软件对天线模型进行仿真优化，并使用优化后的结构参数将天线制作实物样品，如图图2 天线实物图图为实测的驻波比随频率变化曲线，在整个测试频段1～1.8 GHz内，天线的驻波比均小于说明该天线具有良好的阻抗匹配特性。
图线轴比随频率变化曲线，在整个测试频段，天线的轴，说明天线的圆极化性能好。
图3 天线实测驻波比随频率变化曲线图4 天线仿真与实测的轴比随频率变化曲线图5为天线仿真与实测的增益随频率变化的曲 2020年11月25日第37卷第22期Telecom Power TechnologyＮｏｖ． ２５，２０２０，Ｖｏｌ． ３７ Ｎｏ． ２２ 林 飞，等：应用于GNSS终端的宽带扁平状双频圆极化微带天线线。
一种新型宽带鞭状套筒天线康文臣;赵智兵【摘要】套筒天线是种常用的线天线形式。
在常规套筒天线结构的基础上，提出一种新型鞭状套筒天线形式，由双层套筒结构构成。
相对常规套筒，该天线具有对地不敏感、半径小、易于鞭状化等优点。
测量结果表明，天线在220～600 MHz 匹配良好，水平方向增益在0 dBi 以上，实测结果与理论分析吻合良好，验证了该结构在工程应用上的有效性。
%The sleeve antenna is a commonly-used wire antenna.Based on conventional sleeve antenna,a novel flagelliform sleeve antenna is proposed,which is composed of double pared with conventional sleeve antenna,this antenna has such advantages as non-sensitivity to ground,small radius,easy to be flagelliform,etc.The measured results show that the antenna has better matching from 220 MHz to 600 MHz and the horizontal gain is above 0 dBi in bandwidth.The experimental results highly accord with that of theo-ritical analysis,and prove the validity of new structure in engineering.【期刊名称】《无线电工程》【年(卷),期】2014(000)003【总页数】4页(P57-60)【关键词】双套筒;鞭状化;水平方向增益【作者】康文臣;赵智兵【作者单位】国营第七一三厂研究所，江西九江332002;国营第七一三厂研究所，江西九江332002【正文语种】中文【中图分类】TN823+.180 引言套筒单极子天线作为一种宽频带天线在现代化通信和遥感系统中已被广泛采用。
宽带天线刘昌坤（201821020827）天线可将自由空间中的电磁波和接收机／发射机中的输入／输出信号相互转换，是无线电系统中不可或缺的部件。
天线理论与技术有着漫长的发展历史，自赫兹教授1886年发明了第一对收发天线的一百多年来，大量新型天线得以问世。
不同类型、频段的天线得到深入的研究并被广泛应用于实际工程中。
近些年无线通信标准的工作频段不断提高以及对于高速率的要求促使了天线朝着宽带的方向发展，这无疑是一项严峻的挑战。
(a)(b)(c)图1-1 同轴线馈电的UWB天线根据馈电方式的不同，超宽带天线可主要分为同轴线馈电、微带线馈电和共面波导馈电三类。
早期的超宽带天线主要采用同轴线馈电的方式，这种天线往往为立体结构，与同轴线外导体连接的是水平地面，而内导体则接在垂直平面内的辐射体上，通过改变辐射单元的形状及其与地面间的位置参数等，可以实现很宽的工作频带。
Seong-Youp Suh等[1]提出了一种辐射单元为液滴形状的超宽带天线，在辐射单元上开两个圆形的孔，进而实现了更好的匹配效果。
Nader Behdad 等[2]在分析扇形耦合环天线的基础上，优化得到如图1-1(a)所示的超宽带天线，减少了辐射单元所用的金属材料。
上述两个天线的辐射单元均为垂直面内的平面结构，要在水平面内获得更好的全向辐射特性，辐射单元则为沿着馈电接头轴向对称的结构[3-4]。
文献[3]中辐射单元为两个圆形的相交面沿轴向旋转而成，地板的形状也由原来的水平面改成圆锥形结构，天线的最高工作频率高达20GHz 以上，并且在水平面内有很好的全向辐射特性。
图1-1(b)所示为文献[4]中提出的一种圆环形结构的贴片天线，在圆环的外侧和地面之间加入四个短路探针以缩小天线的高度。
实现天线体积小型化的另一种方法如图1-1(c)所示[5]，对辐射单元进行弯折的同时加入短路探针，使其达到接近平面的结构，在保持超宽频带工作的前提下，进一步减小了天线的体积。
平面小型化三频微带天线王公晗;冯全源【摘要】针对多频天线结构复杂,天线尺寸较大,设计了一款紧凑型结构的三频单极性微带贴片天线。
该天线的辐射单元由双C型结构和加载倒L型结构构成,利用低频段的高次模,从而产生天线的高频段。
该方法可以有效实现多频特性,并能够有效地减小天线尺寸。
天线尺寸仅为20×31×1.6 mm3。
实测频段为2.40~2.50 GHz,3.17~3.90 GHz,4.67~5.83 GHz。
该天线具有体积小,结构简单,辐射特性良好的优点,实现了对3.5/5.5 GHz WIMAX频段和2.4/5.2/5.8 GHz WLAN频段的全覆盖,能够很好的适用于无线通信系统的应用。
%This paper proposed a compact structure of tri-band microstrip antenna in order to solve the problems complex structure and the larger size of multi-frequency antenna.The antenna was composed by the two C-ring structures with a pair of inverted L-shaped stubs.Besides,this design utilized high-order mode to generate high frequency.This method could effectively achieve multi-frequency characteristics and reduced the antenna size. The experimental results showed that the antenna had the impedance bandwidths of 100MHz (2.40-2.50 GHz), 730MHz (3.17-3.90 GHz)and 1160 MHz (4.67-5.83 GHz),which could cover both WLAN in the 2.4/5.2/5.8 GHz bands and WIMAX in the 3.5/5.5 GHz bands.【期刊名称】《探测与控制学报》【年(卷),期】2014(000)005【总页数】4页(P64-67)【关键词】天线;三频;高次模;小型化【作者】王公晗;冯全源【作者单位】西南交通大学信息科学与技术学院，四川成都 610031;西南交通大学信息科学与技术学院，四川成都 610031【正文语种】中文【中图分类】TN8210 引言近年来，随着无线通信的迅速发展，各类天线的发展也受到越来越多的关注。
Dual-Feed Dual-Polarized Patch Antenna With Low Cross Polarization and High IsolationChow-Yen-Desmond Sim,Chun-Chuan Chang,and Jeen-Sheen Row Abstract—A dual-feed dual-polarized microstrip antenna with low cross polarization and high isolation is experimentally studied.Two different feed mechanisms are designed to excite a dual orthogonal linearly-polarized mode from a single radiating patch.One of the two modes is excited by an aperture-coupled feed,which is comprised of a compact resonant an-nular-ring slot and a T-shaped microstrip feed line;while the other is ex-cited by a pair of meandering strips with a180phase differences.Both the linearly-polarized modes are designed to operate at2400MHz frequency band,and from the measured results,it is found that the isolation between the two feeding ports is less than40dB across a10dB-input-impedance bandwidth of14%.In addition,low cross polarization is observed from the radiation patterns of the two modes,especially at the broadside direction. Simulation analyses are also carried out to support the measured results.Index Terms—Dual feed,dual polarization,patch antennas.I.I NTRODUCTIONDual-polarized antennas have been widely applied into present wire-less communication systems.Besides the ability to improve the overall system performance by means of polarization diversity,they are also able to provide double transmission channels in a frequency-reuse com-munication system.For microstrip patch antenna,dual linearly-polar-ized operation can be easily realized by using a pair of probe feeds to respectively excite two orthogonal fundamental modes from a single radiating patch;however,some undesirable higher-order modes would also be stimulated at the same time.When a thicker substrate is em-ployed for a microstrip antenna,which is a common method to obtain a broad impedance bandwidth,the transverse currents of the higher-order modes,as well as the currents on the feeding probes may cause obvious cross-polarization radiations[1],[2],and the discrimination between the two linear polarizations is thus degraded.Several methods to re-duce the cross polarization have been demonstrated[2]–[6],and it is found that the higher-order modes can be suppressed when the antenna is symmetrically excited by a dual-feed system with a phase difference of180 ,and if a meandering probe is used instead of a straight probe, the probe radiation can be further reduced.Once the cross polarization at each feeding port of the dual-feed dual-polarized microstrip antenna is suppressed,the isolation level between the two input ports will cer-tainly be improved.Furthermore,the isolation can be further enhanced by employing a two hybrid input ports structure.For example,when both the orthogonal modes are symmetrically excited by meandering probes with phase differences,the isolation level obtained is around 30dB[7],and if an isolation of up to40dB is required,a dual-hy-brid-feed structure(a combination of probe and slot design)is recom-mended[8],however,excess back radiation would be produced by the slot feed,and with thicker antenna substrate,more back radiation will be incurred.Manuscript received April18,2008;revised March23,2009.First published July28,2009;current version published October07,2009.C.-Y.-D.Sim is with the Department of Electrical Engineering,Feng Chia University,Taichung,Taiwan40724,R.O.C.C.-C.Chang and J.-S.Row are with the Department of Electrical Engineering, National Changhua University of Education,Chang-Hua,Taiwan500,R.O.C. Color versions of one or more of thefigures in this communication are avail-able online at .Digital Object Identifier10.1109/TAP.2009.2028702Fig.1.Geometry of the proposed dual-feed dual-polarized microstrip antenna.Therefore,since an isolation level of more than40dB for the designof a dual-feed dual polarized hybrid antenna with low back radiationis not commonly reported elsewhere in the open literature,hence,in this communication,a dual-polarized microstrip antenna designis proposed and studied.In this design,the dual orthogonal linearpolarizations are generated by a new dual-hybrid-feed structure com-prises of a pair of meandering strips with a phase difference of180 ,and a full-wavelength annular-ring coupling slot.Both the feedingmechanisms can symmetrically excite their respective linearly-po-larized modes,thus achieving the characteristics such as low crosspolarization(<23dB)and high isolation for up to40dB.To reducethe back radiation caused by the resonant coupling slot as observed in[8],a novel method of loading additional slots evenly distributed alongthe circumference of the annular-ring slot is introduced.By doing so,the coupling strength between the coupling slot and radiating patchcan be enhanced,and the required size of the annular-ring slot is thusreduced.Note that the loaded slots are analogous to the H-shaped[9]and dog-bone slots[10]designs,which are evolved from a linear slotdesign.For the proposed antenna structure,since both feeding portsare operating at around2.4GHz with an impedance bandwidth ofaround14%,thus it is suitable for WLAN/Bluetooth application infixor mobile wireless base stations.Both experimental and simulationresults are presented and discussed.II.A NTENNA S TRUCTURE AND A NALYSISFig.1presents the configuration of the proposed dual-port dual-po-larized microstrip antenna.The feeding networks of the two input portsare printed on one side of an FR4substrate(thickness1.6mm and per-mittivity4.4),and on the other side is a ground plane embedded with acoupling aperture.The rectangular radiating patch supported by a pairof meandering strips(8mm width)is located10mm above the groundplane.Both the meandering strips are screwed to the bottom of the ra-diating patch,and the positions of the two plastic screws are on y-axisand symmetrical with respect to x-axis.The feeding network of Port1 0018-926X/$26.00©2009IEEETABLE IP ARAMETRIC S TUDIES OF THE P ROPOSED A NTENNA W ITH R ESPECT TO I TS R ESONANCE F REQUENCY (f )ANDI NPUT I MPEDANCE (SWR )AS A F UNCTION OF G EOMETRICAL P ARAMETERS .(SV S MALL V ARIATION)consists of a Wilkinson power divider and a half-wavelength delay line,so that the signals in the two meandering strips will have equal ampli-tudes and 180 phase differences.Consequently,a y -directed polariza-tion mode can be symmetrically excited through Port 1,and the probe radiation can be minimized.As for Port 2,its feeding network is com-prised of a T-shaped coupling strip and an impedance transformer.The feeding network and the coupling aperture in the ground plane form an aperture-coupled feeding mechanism to excite an x -directed polariza-tion mode.The coupling aperture is an integration of an annular-ring slot and six H-shaped slots.When the annular-ring slot is resonating at its fundamental mode,the electric-field distributions in the upper and lower semi-rings will have the same amplitude but with 180 out of phase.This suggest that the single resonant annular-ring slot can be re-garded as a dual-slot feed system with a 180 phase shift,that is able to suppress the undesirable higher-order modes appear at Port 2.As for the H-shaped slots,they are used to reduce the required size of the res-onant coupling slot for a fixed substrate thickness.From the simulated results,after integrating the six H-shaped slots,the mean radius of the annular-ring slot is decreased by 38%.It has to be mentioned that be-sides the mean radius of the annular-ring slot,the tuning of parameter l s is also vital in obtaining a low cross polarization level with respect to the x -directed polarization.Note that l s is the length of the T-shaped coupling strip being protruded out from underneath the annular ring slot.III.R ESULTS AND D ISCUSSIONThe optimum performance of the proposed antenna is achieved by the parametric studies carried out by HFSS,and its detailed optimum dimensions are as depicted in Fig.1.The design guideline is presented in Table I,whereby the sensitivity analysis of the proposed antenna with respect to its resonance frequency (f )and input impedance (SWR )as a function of each geometrical parameter are presented;for example,widening or reducing the width of the meander strip width W m will only results in a small variation in resonance frequency and impedance bandwidth.As for the tuning of parameter l s ,it is achieved by reducing or increasing both the length of the vertical T-shaped coupling strip as shown in Fig.1.From the parametric studies in Table I,an increase in l s will results in a reduction of resonance frequency and impedancebandwidth.Fig.2.Return loss against frequency for Port 1and Port 2of the constructed prototype.The return loss measured at the two feeding ports of the proto-type is presented in Fig.2,along with the simulated results.Note that the differences between them,especially at Port 1,is due to the imperfect construction (handmade)of the 2meander strips that are supporting the radiating copper plate,which shows a much smaller loci loop and slight phase deviation in the smith chart.Therefore,the supposed double resonances as shown in Port 1(simulated)are being combined as a single resonance for the constructed prototype as shown in the measured value.To further proven that the discrepancies in Fig.2will not seriously affect the performances,the variation of impedance against frequencies for Port 1are presented,which shows a well validated simulated and measured resistance/reactance values in Fig.3.Furthermore,the measured phase accuracy of the balanced feed lines (Port 1)in Fig.4also shows a relative phase shift of ex-actly 180degree at around 2440MHz.From the obtained results in Figs.2and 3,good excitation at around 2400MHz are exhibited by both the feeding ports,and the impedance bandwidth (also referred to as 10dB return loss)measured at Port 1and Port 2,with respect to the center frequency measured at 2450and 2420MHz,respectively,is 14.2and 14.5%.Although these two center frequencies are very close to each other,it is noteworthy that the radiating copper patch is not a square shape,which in fact has an aspect ratio of 1.13.This implies that the required resonant length of the x -polarized mode forFig.3.Measured and simulated impedance of Port1.Fig.4.Measured and simulated isolation level,and measured relative phase shift against frequency of the constructed prototype.this case is obviously different from that of the y -polarized mode,es-pecially when these two modes are resonating at the same frequency.Therefore,the use of a rectangular radiating patch is helpful in re-ducing the coupling between the dual orthogonal modes.The isolation level between the two feeding ports of the prototype is shown in Fig.4,and an isolation of more than 40dB over the entire impedance bandwidths is demonstrated.Although there is a difference between the measured and simulated results after 2500MHz,this could be due to the very low (isolation)signal level measured above 40dB,in which this case,the VNA (or the cables)used might not respond ac-curately at such level.Furthermore,we have to consider that the con-structed prototype is imperfect as compared to the simulated one,not to mention also that such isolation level (>40dB)is not commonly reported in the open literature for the design of a dual-feed dual-polar-ized microstrip antenna.Therefore,since the two feeding ports are well decoupled,low cross-polarization radiation can be expected at each feeding port.The radiation patterns measured at 2400MHz for Port 1and Port 2are plotted in Fig.5.Good broadside radiation patterns with low cross polarizations are observed in the two principal planes.For Port 1,the cross polarization level is 027dB in the E-plane and 023dB in the H-plane.As for Port 2,the cross polarization level is 030dB in the E-plane and 025dB in the H-plane.From these results,it is interesting to learn that the peak gains measured at both ports are almost the same at around 7.4dBi,and in addition,the front-to-back (F/B)ratios for Port 1and Port 2are 18and 13dB,respectively.As compared to the case without the H-shaped loading slots,the measured F/B ratio for Port 2will only be 8dB.Fig.5.Measured radiation patterns at 2400MHz for Port 1and Port 2of the constructed prototype.(a)E-plane for Port 1,(b)H-plane for Port 1,(c)E-plane for Port 2,(d)H-plane for Port 2.Although the propagation environment will highly influence the discrimination between the orthogonal polarizations,it is still worth pointing out that for this proposed dual-polarized antenna,more atten-tion should be focused on the cross polarization level at the direction with maximum gain,rather than other directions.By observing each radiation pattern in Fig.5,it is obvious that all of the cross-polarization fields have a dip point on the z -axis,which also exhibits a very low on-axis cross polarization level at both the feeding ports.Therefore,a good discrimination between the two orthogonal polarizations at the direction with maximum gain is achieved.Note that the occurrence of this dip point is due to the transverse currents induced by the higher-order modes on the radiating patch that are symmetrical,and their radiations is nearly cancelled at the broadside direction.Fig.6presented the on-axis cross polarization level of both feeding ports against frequency within the impedance bandwidth.For both Port 1and Port 2,their minimum values at the on-axis cross polarization level occur at around 2400MHz,whereby the delay line is just half wavelength long and the annular-ring slot is resonant at this frequency.In Fig.6,a slight disagreement between the simulated and measured value for Port 2is partly due to the imperfect constructed prototype,and it is also suspected that such low cross polarization signal level below 040dB (Port 2-simulated)have reached the limitation of our measurement system.Nevertheless,similarity in the trend for both measured and simulated cross polarization level is presented.From the measured results in Fig.6,it is also observed that the operating frequency for both feeding ports are located within an on-axis cross polarization level of less than 030dB,and their overlapped fre-quencies are from 2380to 2460MHz.Note that these overlapped frequencies are very close to the WLAN/Bluetooth operating band from 2400to 2484MHz,and the simulated efficiency within the WLAN/Bluetooth band for Port 1is between 75.6to 86.4%,while Port 2shows between 81.5to 84.5%.A lower efficiency level for Port 1could be due to the power dissipation of the chip resistor.To clarifyFig.6.Variations of the on-axis cross polarization level against frequency for Port 1and Port 2of the constructedprototype.Fig.7.Simulated current distribution at 2400MHz for Port 1(left)and Port 2(right).if the orthogonal radiations are emitted mainly from the copper patch or from the ground plane,the current distribution for both feeding ports at 2400MHz are presented in Fig.7.Ignoring the feeding line of both ports and the coupled radiation found at the centre of the ground plane,the radiating copper patch is the main resonator for the orthogonal radiations at boresight direction.Since the proposed antenna is able to achieve excellent isolation and cross-polarization level with good peak gains,it is therefore also pos-sible for antenna array design and CP radiation.As for the array de-sign,the downside includes complication in design (complex feeding network)which leads to difficulty in fabrication.Although good CP ra-diation can be emitted from the proposed antennas,an additional (or ex-ternal)quadrature feeding network with balanced feed line is required to insert into the input ports,which will further complicate the design of the proposed antenna.Hence,additional research on CP performances will be carried out in the future.IV .C ONCLUSIONA dual-feed dual-polarized microstrip antenna has been presented in this communication.Two linearly-polarized modes of the antenna are excited by a dual-hybrid-feed structure with a combination of probe and slot design.To reduce the radiation from the feeding structure,me-andering strips are used instead of a conventional straight probe,and the coupling slot is embedded with six H-shaped slots.To suppress theundesirable higher-order modes,each linearly-polarized mode is pro-duced by the technology of a dual-feed system with 180 phase differ-ences.From the experimental results,it is found that the isolation level between the two feeding ports is less than 40dB within an impedance bandwidth of around 14%.The measured far-field pattern results also demonstrated that the two linear polarization radiations have a sym-metrical main beam,and their cross polarization levels in both E-and H-planes are less than 023dB.In addition,a very low on-axis cross polarization level is observed,and therefore the two linear polarization senses can be easily distinguished from each other at the direction with maximum gain.R EFERENCES[1]T.Huynh,K.F.Lee,and R.Q.Lee,“Crosspolarisation characteristicsof rectangular patch antennas,”Electron.Lett.,vol.24,pp.463–464,Apr.14,1988.[2]Z.N.Chen and M.Y.W.Chia,“Experimental study on radiation per-formance of probe-fed suspended plate antennas,”IEEE Trans.An-tennas Propag.,vol.51,pp.1964–1971,Aug.2003.[3]P.Li,i,K.M.Luk,and u,“A wideband patch antennawith cross-polarization suppression,”IEEE Antennas Wireless Propag.Lett.,vol.3,pp.211–214,2004.[4]A.Petosa,A.Ittipiboon,and N.Gagnon,“Suppression of unwantedprobe radiation in wideband probe-fed microstrip patches,”Electron.Lett.,vol.35,pp.355–357,Mar.4,1999.[5]X.Y.Zhang,Q.Xue,B.J.Hu,and S.L.Xie,“A wideband antennawith dual printed L-probes for cross-polarization suppression,”IEEE Antennas Wireless Propag.Lett.,vol.5,pp.388–390,2006.[6]C.H.K.Chin,Q.Xue,H.Wong,and X.Y.Zhang,“Broadband patchantenna with low cross-polarisation,”Electron.Lett.,vol.43,pp.137–138,Feb.1,2007.[7]i and K.M.Luk,“Dual polarized patch antenna fed by mean-dering probes,”IEEE Trans.Antennas Propag.,vol.55,pp.2625–2627,Sept.2007.[8]T.W.Chiou and K.L.Wong,“Broad-band dual-polarized single mi-crostrip patch antenna with high isolation and low cross polarization,”IEEE Trans.Antennas Propag.,vol.50,pp.399–401,Mar.2002.[9]H.S.Shin and N.Kim,“Wideband and high-gain one-patch microstripantenna coupled with H-shaped aperture,”Electron.Lett.,vol.38,pp.1072–1073,Sep.12,2002.[10]D.M.Pozar and S.D.Targonski,“Improved coupling for aperture cou-pled microstrip 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Progress In Electromagnetics Research C,Vol.16,147–160,2010A MEANDERED LOOP ANTENNA FOR LTE/WW AN OPERATIONS IN A SMART PHONEC.-W.Chiu and C.-H.ChangDepartment of Electronic EngineeringNational Ilan UniversityIlan260,TaiwanY.-J.ChiDepartment of Electrical EngineeringNational Chiao-Tung UniversityHsinchu350,TaiwanAbstract—This paper presents a multiband meandered loop antenna for smart phone applications.The proposed antenna which features a meandered folded-loop generates two resonance modes in the LTE/GSM bands.The current distributions of the excited resonance modes are analyzed to investigate the mode characteristic.By using a capacitively coupled feed on the backplane,the impedance bandwidth is broadened to cover LTE/WWAN bands.The simulation performed in this research used a high frequency structure simulator (HFSS)to optimally design the antenna,and a practical structure was constructed for the test.Details of the various antenna parameters are presented and discussed in this paper.1.INTRODUCTIONLTE(Long Term Evolution)is a new high-performance air interface standard for cellular mobile communication systems.It is the last step toward the4th generation(4G)of radio technologies designed to increase the capacity and speed of mobile telephone networks.LTE provides ultra-broadband speeds for mega multimedia applications by using a high performance antenna[1].The frequency spectrum allocated for LTE applications ranges from600MHz to3GHz,and Received25July2010,Accepted13September2010,Scheduled26September2010 Corresponding author:C.-W.Chiu(alexchiu@).148Chiu,Chang,and Chi the LTE band12–14covers from698MHz to798MHz.For the incorporation of the LTE band with the existing cellular phone,the operating bandwidth of a handset should cover from698MHz to 960MHz and from1710to2690MHz[1].In order to include LTE700/LTE2300/LTE2500,Wong and Chen recently proposed a small-size printed loop antenna integrated with two stacked coupled-fed shorted strip monopoles for multiband operation in a mobile phone[2].Some published papers proposed a multi-input and multi-output(MIMO)antenna configuration for LTE handset application in order to deliver ultra-broadband speeds for mega multimedia applications[3,4].Since the spectrum for the LTE700 system is allocated at700MHz bands,the operating wavelength is longer than400mm.The ground plane size of a typical handset(say 100mm)is only about a quarter-wavelength.As a result,the chassis of a traditional monopole or PIFA in the handsets is a resonator.The resonating currents are spread out over the system’s ground plane[5–8].To avoid degrading of the antenna performance,a loop antenna is a good candidate for a LTE system.The performance of the loop antenna is less dependent on the ground plane,thus it is suitable for a LTE antenna design[9].The main current distribution of the loop antenna is limited in the closed loop pattern and feed port.Therefore, handheld and head proximity influences are reduced since the current of the loop antenna on the ground plane is less than that of the PIFA or monopole[10–14].In this paper,a compact internal antenna which operates in the LTE/GSM and PCS/UMTS/WLAN/LTE2300bands is proposed.The antenna is a kind of folded loop which is fed by a back-coupling element connected to a microstrip transmission line[15].A parallel monopole-like mode and a combination mode formed by loop pattern and device chassis are excited to cover the bandwidth from698MHz to960MHz. This design is easily to be implemented in a double-sided printed circuit board.The occupied geometrical space of the antenna structure measures only60mm×10mm×6.5mm.The proposed antenna design is described in Section2,and a parameter study on analyzing the effect of some critical parameters is also presented.Experimental results of the proposed prototype are presented and discussed in Section3.2.ANTENNA DESIGN AND ANALYSIS2.1.Antenna DesignFigure1(a)shows the three-dimensional configuration of the proposed antenna.The antenna is mounted on a0.8-mm thick FR4substrate with a relative permittivity of4.4and a loss tangent of0.02.TheProgress In Electromagnetics Research C,Vol.16,201014960110unit: mm6.550Ω microstrip feeding line on the back sideground plane 60x100CDFYZ XH27.527.5strip line A(a)(b)unit:mmn=22m =17.7d=650 Ω microstrip feeding line29.2529.25 Lg=1001ground plane on the back side 100 x 60top edge of ground plane (on the back side )coupling elementtuning stub(c)Figure 1.Geometry of the proposed antenna.(a)3D view,(b)plan view of the front-side,and (c)plan view of the back-side.antenna system consists of a folded loop strip and a capacitively-coupled feeding line.The loop pattern is meandered and folded to increase the electrical length but at the same time reduce the size it occupies.The total length of the folded and meandered loop strip from C to D is about 286mm,as shown in Fig.1(b).To excite a new resonant mode by coupling,a strip line A is arranged directly above the coupling element on the back plane.To increase electric length and support the meandered loop,a tuning strip line B is inserted into the loop.On the same side of the substrate,a copper plate which is 60mm wide and 100mm long is printed to act as the system ground plane of a smart phone.To broaden the impedance bandwidth,an inverted L-shape coupling element with a matching stub,as shown in Fig.1(c),is placed on the back side and connected to the feed port.The150Chiu,Chang,and Chicoupling feed scheme generates two resonance modes in the LTE/GSM band.The coupling element on the backplane is connected to a 50ohm microstrip line with a strip width of 1.5mm.Figure 2(a)shows the simulated reflection coefficient for the folded loop antenna fed by the capacitively-coupled microstrip line compared with the one fed directly by a coaxial cable at point C shown in Fig.1(b).Basically,a parallel monopole-like mode,which its current path is from point D,through E,G,to H,is excited on the loop pattern by the unbalance-fed scheme.If the antenna is fed directly,the monopole-like mode is generated on the two arms from D to H and C to H in the lower GSM band.The field of the antenna system with the direct feeding scheme is inductively coupled to the ground plane.Since the maximum magnetic field is located at the center of the ground plane,not near the rim of the conductor plane,the antenna system cannot generate a chassis wavemode.However,two resonating modes are excited when the antenna is capacitively fed on the backside of the substrate because the maximum electric field of the chassis mode is close to the rim of the ground plane.Hence,the PCB resonance mode is easily excited by capacitive coupling.As a result,the impedance bandwidth could be broadened to cover the LTE/GSM850/GSB900bands.Figure 2(b)shows the input impedance when the antenna is fed by the two different feed schemes.The reactance of the impedance in the low band is inductive when the antenna is fed directly by a cable at point C.This inductive reactance has a high rate of change with respect to frequency so as to limit the bandwidth;therefore,the PCB chassis mode cannot be generated.Consider the loop antenna has a back-coupled feed port.The inductive reactance is compensated by theFrequency (GHz)R e f l e c t i o n C o e f f i c i e n t (d B )direct feedingcapacitively-coupling-6dBR e [Z i n ](Ω)I m [Zi n](Ω)Frequency (GHz)0.511.522.53-30-25-20-15-10-50200400600800(a)(b)Figure 2.Simulated results of (a)the reflection coefficient and (b)the input impedance with different feed schemes.Progress In Electromagnetics Research C,Vol.16,2010151Frequency (GHz)R e f l e c t i o n C o e f f i c i e n t (d B )Figure 3.Simulated results of the reflection coefficient with strip lines A and B.capacitively coupled element to form a self-complimentary impedance so as to widen the impedance bandwidth (red line).The input signal launched from the inverted-L-shape feed line capacitively couples to strip line A.Strip line A helps to excite a new resonant mode.Fig.3shows the reflection coefficient for the antenna with and without strip lines A and B.A new loop mode is excited when strip line A is inserted.The new chassis-handset combination,which is generated from point C’,through F,E,G to H and C’,is supported by the coupled structure with strip line A,as Fig.1(b)shows.On the other hand,a parasitic line B is inserted into the loop to increase the electrical length of the lowest resonance frequency and support the folded loop.When inserting the strip lines A and B,this study finds that the bandwidth is broadened because two modes are generated in the low GSM band.2.2.Current DistributionThe surface current distribution on the conductor strip confirms the mode characteristic.Fig.4shows the vector current distributions at 720and 990MHz.The current behavior demonstrates that the monopole-like mode creates the resonance at 720MHz and the antenna-chassis combination mode at 990MHz,respectively.These results are simulated by a commercial electromagnetic simulation tool,HFSS.The strip ends of the loop antenna structure symmetrically terminate to the ground plane.The currents on both sides of the loop are of the same phase at 720MHz with respect to the central feed line,as Fig.4(a)shows.Thus,the antenna behaves like two parallel folded monopoles at 720MHz.The quarter wavelength at 720MHz is calculated to be around 104mm but the meander length of one arm,as shown in152Chiu,Chang,and Chi(a)(b)Figure4.Simulated vector current distributions and surface current densities.(a)720MHz,(b)990MHz.Progress In Electromagnetics Research C,Vol.16,2010153Fig.1(b),is around 141.5mm.The physical length of the wavelength of the monopole-like mode is longer than the electrical length due to fold and meander.The monopole-like mode can also be verified by the radiation patterns discussed in Section 3,where E Θin the x -y plane is basically omni-directional.The current behavior in Fig.4(b)shows that the antenna-chassis combination mode is excited at 990MHz.Since the proposed antenna has an unbalanced feeding scheme,the currents are generated on the groundplane,and therefore,the ground plane is operated as a part of a radiator.Since the ground length is smaller than half-wave length below 1GHz,the resonance mode at 990MHz uses the printed circuit board of the mobile terminal as part of the antenna.The characteristic wave modes of the PCB conductor have contribution on the combined radiation behavior of the PCB and the loop element.The radiation mode is characterized as the combination of the handset antenna and the PCB conductor [16].The current distribution at 990MHz shows that it doesn’t form complete resonance current path in the loop pattern.The antenna element dominantly acts as a coupling element at 990MHz.Therefore,the antenna system can be modeled as a dual-resonant circuit [16].The radiation mode due to the groundplane resonance enhances the bandwidth when the antenna system is properly designed [7].Fig.5shows the effect of ground length Lg on the antenna performance,and observations showed larger effects on the resonance modes in the low band.The longer the length,the greater the bandwidth [7,16].Frequency (GHz)R e f l e c t i o n C o e f f i c i e n t (d B )Figure 5.Simulated results as a function of length L g (system ground planelength).-5Frequency (GHz)R e f l e c t i o n C o e f f i c i e n t (d B )Figure 6.Simulated results as a function of length m .154Chiu,Chang,and ChiFrequency (GHz)R e f l e c t i o n C o e f f i c i e n t (d B )Figure 7.Simulated results as a function of length n of the tuning stub.Frequency (GHz)R e f l e c t i o n C o e f f i c i e n t (d B )Figure 8.Simulated results as a function of distance d of the tuning stub.2.3.Parametric AnalysisTo obtain better impedance matching,an inverted L-shape coupling element with a tuning stub is connected to the feeding microstrip line on the back plane.The design of the coupling element is a critical factor.This research conducted a parametric study on the length m and the stub length n shown in Fig.1(c).When the parameter m varies between 14.7to 24.7mm or the parameter n varies from 19to 25mm,the rest of the dimensions of the antenna remain the same as shown in Fig.1.Fig.6shows the simulation reflection coefficient as a function of length m .Length m plays an important role in exciting the second mode in the low band.The variation in length has a substantial influence on the impedance matching in the design bands.Fig.7shows the simulation reflection coefficient as a function of stub length n .The findings show that the stub length n has a larger effect in the higher band.When the length is tuned to about 22mm,the bandwidths in the low band and the higher band achieve the bandwidth requirement of −6dB.Fig.8shows the simulation reflection coefficient as a function of d which is the distance from the ground plane edge to the tuning stub shown in Fig.1(c).When distance d is fixed at 6mm,the impedances are well matched in the higher band.3.RESULTS OF THE PROPOSED ANTENNAThe proposed antenna was constructed for testing purposes.The measurements were performed using an Agilent E5071B network analyzer.Fig.9shows the measured and simulated reflectionProgress In Electromagnetics Research C,Vol.16,2010155Frequency (GHz)R e f l e c t i o n C o e f f i c i e n t (d B )Figure 9.Measured results compared with simulation results.coefficient of the proposed antenna.The simulated results were obtained using Ansoft HFSS.There was a good agreement between the measurement and simulation below 1.8GHz.The finding shows that the achieved bandwidth covers the LTE/GSM850/GSM900band and PCS/UMTS/WLAN/LTE2300.In the low band,the achieved bandwidth with reflection coefficient better than −6dB was 423MHz (675–1098MHz)and in the high band it was 870MHz (1760–2630MHz).Figure 10shows the measured and simulated radiation patterns of the proposed antenna on the XY plane.The radiation-pattern and gain measurements were performed in the anechoic chamber of SGS pany in Taiwan.The radiation patterns E Θare nearly omni-directional in the xy -plane.These features are desirable characteristics for mobile phones.Fig.11shows the measured peak gains and radiation efficiencies.The peak gains and the radiation efficiencies were measured using the ETS-Lindgren model AMS-8500antenna measurement system and the 3164-08open boundary quad-ridged horn antenna,respectively.The measured antenna gain over the low band varied from around 1.5–3.0dBi and the radiation efficiency was higher than 50%.The measured antenna gain for the higher band varied around 1–4.2dBi and the radiation efficiency was larger than 50%,ranging from 1.8to 2.5GHz.Since the connecting semi-rigid coaxial cable (about 30cm)between the DUT and the testing system of SGS is not wrapped by a choke or a ferrite bead,the connecting cable used in the measurement acts as a radiator in the lower band (below 1GHz)due to the currents traveling on its outer face [17].The additional radiation from the outer conductor of the cable has some impact on the gain.Therefore,the measured gain is larger than that156Chiu,Chang,and Chiof ordinary monopoles in the lower band.The specific absorption rate (SAR)of the proposed antenna was studied using the FEKO simulation software [18].Fig.12shows the simulation model with the antenna placed at the cheek position near the phantom.The phantom of the head model used in SAR computation is a specific anthropomorphic mannequin (SAM)defined by the IEEE Standards Coordinating Committee 34[19].The separation distance between the system ground plane and the earpiece of the SAR phantom head is 5mm.The tilted angle between the center2045MHz2350MHzX-Y planeY270180270180(d)(e)Figure 10.Measured and simulated radiation patterns at 745MHz,850MHz,925MHz,2.045GHz,and 2.35GHz,respectively.line of the printed circuit board and the vertical line of the phantom is 63◦.The antenna is placed at the top edge of the system ground plane or at the bottom position (rotating 180◦).The testing power is 24dBm (0.25W)for the GSM850/GSM900/UMTS bands,while the testing power is 20.8dBm (0.121W)for the DCS/PCS bands.Table 1lists the simulated 1-g average SARs at the transmitting frequencies of 836.6,914.8,1747,1880,and 1950MHz.The finding shows that the SAR at the bottom position is lower than that at the top position owing to the larger distance to the cheek.It seems that placing the antenna at the bottom edge is more promising for practical mobile phone applications.Table 1.Simulated 1-g average SAR(W/kg)at GSM850/GSM900-/DCS/PCS/UMTS bands.Band GSM850GSM900DCS PCS UMTS Frequency (MHz)f =836.6f =914.8f =1747f =1880f =1950At the top (W/kg) 1.866 1.887 1.734 1.712 3.281At the bottom(W/kg)1.1771.1490.7390.8911.458Frequency (GHz)A n t e n n a G a i n (dB i )R a d i a t i o n E f f i c i e n c y (%)A n t e n n a G a i n (dB i )R a d i a t i o n E f f i c i e n c y (%)Frequency (GHz)(a)(b)Figure 11.Measured peak gain and radiation efficiency,(a)in the low band and (b)in the higher band.Figure12.Simulation model with the proposed antenna for the SAR analysis.4.CONCLUSIONThis paper proposes a folded and meandered loop antenna for smart phone applications.By using a capacitively coupled feed on the back plane,two resonance modes excited on the meandered loop pattern and the ground plane have been demonstrated to achieve wideband in the lower band.The parametric study on the coupling element was performed to optimally design the folded loop antenna. The measured results on the constructed antenna were presented to validate the proposed design.The achieved bandwidth ranges from675to1098MHz and1760to2630MHz,and the measured results indicate that they cover LTE and WWAN bands.Since the proposed loop antenna is designed to include the new emerging LTE700/LTE2300/LTE2500bands,it is very suitable for the use in the4G smart phone.ACKNOWLEDGMENTWe are grateful to the National Center for High-performance Computing for the HFSS computer time and use of facilities.Also, the authors would like to thank Mr.Yu-Chou Chuang and Mr.Cheng-Chang Chen,Bureau of Standards,Metrology and Inspection, M.O.E.A,Taiwan,for their help in the SAR simulation using the FEKO simulation tool.REFERENCES1.Sesia,S.,I.Toufik,and M.Baker,LTE—The UMTS Long TermEvolution:From Theory to Practice,Wiley,Chichester,UK,2009.2.Wong,K.L.and W.Y.Chen,“Small-size printed loo-typeantenna integrated with two stacked coupled-fed shorted strip monopoles for eight-band LTE/GSM/UMTS operation in the mobile,”Microwave and Optical Technology Letters,Vol.52,No.7, 1471–1476,Jul.2010.3.Chaudhury,S.K.,H.J.Chaloupka,and A.Ziroff,“NovelMIMO antennas for mobile terminal,”Proceedings of the1st European Wireless Technology Conference,330–333,Amsterdam, Netherlands,Oct.2008.4.Bhatti,R.A.,S.Yi,and S.O.Park,“Compact antenna arraywith port decoupling for LTE-standardized mobile phones,”IEEE Antennas and Wireless Propagation Letters,Vol.8,1430–1433, 2009.5.Abedin,M.F.and M.Ali,“Modifying the ground plane and itseffect on planar inverted-F antennas(PIFAs)for mobile phone handsets,”IEEE Antennas and Wireless Propagation Letters, Vol.2,226–229,2003.6.Anguera,J.,I.Sanz,A.Sanz,A.Condes,D.Gala,C.Puente,and J.Soler,“Enhancing the performance of handset antennas by means of groundplane design,”IEEE International Workshop on Antenna Technology:Small Antennas and Novel Metamaterials (IWAT),29–32,New York,USA,Mar.2006.7.Cabedo, A.,J.Anguera, C.Picher,M.Rib´o,and C.Puente,“Multi-band handset antenna combining PIFA,slots,and ground plane modes,”IEEE Transactions on Antennas and Propagation, Vol.57,No.9,2526–2533,Sep.2009.8.Picher,C.,J.Anguera,A.Cabedo,C.Puente,and S.Kahng,“Multiband handset antenna using slots on the ground plane: Considerations to facilitate the integration of the feeding transmission line,”Progress In Electromagnetics Research C, Vol.7,95–109,2009.9.Lin,C.I.and K.L.Wong,“Internal meandered loop antenna forGSM/DCS/PCS multiband operation in a mobile phone with the user’s hand,”Microwave and Optical Technology Letters,Vol.49, No.4,759–765,Apr.2007.10.Chi,Y.W.and K.L.Wong,“Internal compact dual-band printedloop antenna for mobile phone application,”IEEE Transactions on Antennas and Propagation,Vol.55,No.5,1457–1462,May2007.11.Wong,K.L.and C.H.Huang,“Printed loop antenna witha perpendicular feed for penta-band mobile phone application,”IEEE Transactions on Antennas and Propagation,Vol.56,No.7, 2138–2141,Jul.2008.12.Chi,Y.W.and K.-L.Wong,“Compact multiband folded loopchip antenna for small-size mobile phone,”IEEE Transactions on Antennas and Propagation,Vol.56,No.12,3797–3803,Dec.2008.13.Chiu,C.W.,C.H.Chang,and Y.J.Chi,“Multiband foldedloop antenna for smart phones,”Progress In Electromagnetics Research,Vol.103,123–136,2010.14.Chiu,C.W.and Y.J.Chi,“Printed loop antenna with a U-shaped tuning element for hepta-band laptop applications,”IEEE Transactions on Antennas and Propagation,Vol.58,No.11, Nov.2010.15.Li,W.Y.and K.L.Wong,“Seven-band surface-mount loopantenna with a capacitively coupled feed for mobile phone application,”Microwave and Optical Technology Letters,Vol.51, No.1,81–88,Jan.2009.16.Vainikainen,P.,J.Ollikainen,O.Kivekas,and K.Kelander,“Resonator-based analysis of the combination of mobile handset antenna and chassis,”IEEE Transactions on Antennas and Propagation,Vol.50,No.10,1433–1444,Oct.2002.17.Chen,Z.N.,N.T.Yang,Y.X.Guo,and M.Y.W.Chia,“An investigation into measurement of handset antennas,”IEEE Trans.Instrumentation and Measurement,Vol.54,No.3,1100–1110,Mar.2005.18.FEKO,EM Software&Systems—S.A.(Pty)Ltd.(EMSS),[Online],available:.19.Beard, B. 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共面波导馈电的超宽带天线设计房泽;吴萍;孙兵兵【摘要】文章设计了一种共面波导馈电的小型平面超宽带天线.天线由树形辐射单元和共面波导构成,体积小,在工作带宽内具有稳定的方向特性.利用电磁仿真软件对影响天线性能的主要参数进行了仿真、分析和优化,得到了天线的理想尺寸.对优化后的超宽带天线进行制作和测试,测试结果显示天线的工作带宽为3～11 GHz.测试结果与仿真结果吻合,从而证明利用共面波导馈电的超宽带天线设计方法的有效性.%A compact coplanar waveguide(CPW)-fed ultra-wideband(UWB) antenna is presented in this paper. The antenna is formed by the tree-like radiation unit and the CPW. The antenna has the advantages of small size and stable radiation pattern in the operating bandwidth. The key parameters affecting the performance of the antenna are simulated, analyzed and optimized by using electromagnetic simulation software. The ideal dimensions of the antenna are obtained, and the optimized antenna is manufactured and tested The result shows the antenna's bandwidth is 3-11 GHz. The experimental result is in agreement with the simulated result, which shows the design of CPW-fed UWB antenna is feasible.【期刊名称】《合肥工业大学学报（自然科学版）》【年(卷),期】2012(035)004【总页数】4页(P493-495,556)【关键词】超宽带天线;共面波导;微带天线;小型化【作者】房泽;吴萍;孙兵兵【作者单位】安徽大学电子信息工程学院,安徽合肥 230039;安徽大学电子信息工程学院,安徽合肥 230039;安徽大学电子信息工程学院,安徽合肥 230039【正文语种】中文【中图分类】TN914自2002年美国联邦通信委员会（FCC）将3.1～10.6GHz频段划为民用频段后，超宽带系统越来越受到人们的关注，许多新出现的超短波通信系统都工作在此频段内［1］。
Revised 5/2/03Any 2.4GHz Wireless Product Including:■Bluetooth ■802.11■ZigBee■Wireless PCMCIA Cards ■Telemetry■Data Collection■Industrial Process Monitoring ■Compact Wireless Products ■External Antenna EliminationAPPLICATIONS■Incredibly Compact SMD Package ■Superior LTCC T echnology ■50ΩCharacteristic Impedance ■Low Loss■Wide Bandwidth■Favorable Linear Polarization ■> Unity Gain■No External Matching Required ■Highly Stable Over Temp.and Humidity■Fully Hand- and Reflow-Assembly Compatible ■Cost-EffectiveFEATURESThe exciting ANT -2.45-CHP is of the one of the world’s smallest, high-performance 2.4 Ghz Chip Antennas.It is ideal for all 2.4GHz applications including Bluetooth, 802.11, home RF , ZigBee and other popular and emerging standards.The antenna uses an advanced multilayer LTCC Technology and a proprietary hybrid spiral element to achieve size and performance characteristics superior to other designs.The incredibly compact SMD package measures a mere 6.5mm (L) x 2.2mm (W) x 1.0mm (H) and is fully compatible with hand- and reflow-attachment processes.The antenna's favorable electrical specifications, stability and cost-effectiveness make it the logical choice for a wide variety of applications.ANT-2.45-CHP-xDESCRIPTIONPHYSICAL DIMENSIONS2.45GHz ULTRA COMPACT CHIP ANTENNA DATA GUIDEActual SizePad Layout-30-20-10180270-40-30-20-10[deg.]dBSPECIFICATIONSCHARACTERISTICSImpedanceRadiation PatternReturn LossPage 2Page 3REFLOW SOLDERING PROFILEFLOW SOLDERING PROFILE230°C200°C150°C235¡150¡SOLDERING CONSIDERATIONSHand SolderingThis antenna is designed for high-volume automated assembly, however, it may be successfully attached by hand assembly techniques.A hand-solder temperature of 225°or lower should be used.Do not exceed a 10 sec.heating time.Reflow Temperature ProfileThe single most critical stage in the automated assembly process is the reflow process.The reflow profile below should be closely followed since excessive temperatures or transport times during reflow will irreparably damage the antennas.Assembly personnel will need to pay careful attention to the oven's profile to insure that it meets the requirements necessary to successfully reflow all components while still meeting the limits mandated by the antennas themselves.Page 4LINX TECHNOLOGIES,INC.575 S.E.ASHLEY PLACE GRANTS PASS,OR 97526Phone:(541) 471-6256FAX:(541) 471-6251U.S.CORPORATE HEADQUARTERS:Linx Technologies is continually striving to improve the quality and function of its products;for this reason, we reserve the right to make changes without notice.The information contained in this Data Sheet is believed to be accurate as of the time of publication.Specifications are based on representative lot samples.Values may vary from lot to lot and are not guaranteed.Linx T echnologies makes no guarantee, warranty, or representation regarding the suitability of any product for use in a specific application.None of these devices is intended for use in applications of a critical nature where the safety of life or property is at risk.The user assumes full liability for the use of product in such applications.Under no conditions will Linx T echnologies be responsible for losses arising from the use or failure of the device in any application, other than the repair, replacement, or refund limited to the original product purchase price.Some devices described in this publication are patented.Under no circumstances shall any user be conveyed any license or right to the use or ownership of these patents.Disclaimer©2002 by Linx Technologies, Inc. The stylized Linx logo, Linx, and “Wireless Made Simple”are the trademarks of Linx Technologies, Inc. Printed in U.S.A.。
