Chaos in the Colpitts oscillator

TL F 7398DS8911 DS8913AM FM TV Sound Up-Conversion Frequency SynthesizerApril 1990DS8911 DS8913AM FM TV Sound Up-Conversion Frequency SynthesizerGeneral DescriptionThe DS8911is a digital Phase-Locked Loop (PLL)frequen-cy synthesizer intended for use as a Local Oscillator (LO)in electronically tuned radios The device is used in conjunc-tion with a serial data controller a loop filter some varactor diodes and several passive elements to provide the local oscillator function for both AM and FM tuningThe conventional superheterodyne AM receiver utilizes a low IF or down conversion tuning approach whereby the IF is chosen to be below the frequencies to be received The DS8911PLL on the other hand utilizes an up-conversion technique in the AM mode whereby the first IF frequency is chosen to be well above the RF frequency range to be tuned This approach eliminates the need for tuned circuits in the AM frontend since the image half IF and other spuri-ous responses occur far beyond the range of frequencies to be tuned Sufficient selectivity and second IF image protec-tion is provided by a crystal filter at the output of the first mixerA significant cost savings can be realized utilizing this up-conversion approach to tuning Removal of the AM tuned circuits eliminates the cost of expensive matched varactor diodes and reduces the amount of labor required for align-ment down from 6adjustments to 2 Additional cost savings are realized because up-conversion enables both the AM and FM bands to be tuned using a single Voltage Controlled Oscillator (VCO)operating between 98and 120MHz (The 2to 1LO tuning range found in conventional AM down con-version radios is reduced to a 10%tuning range 9 94MHz to 11 02MHz)Up-conversion AM tuning is accomplished by first dividing the VCO signal down by a modulus 10to obtain the LO signal This LO in turn is mixed on chip with the RF signal to obtain a first IF at the MIXER output pins This first IF after crystal filtering is mixed (externally)with a reference fre-quency provided by the PLL to obtain a 450kHz second IFfrequency The DS8911derives the 450kHz second IF by mixing an 11 55MHz first IF with a 12 00MHz reference frequencyFM and WB (weather band)tuning is done using the con-ventional down conversion approach Here the VCO signal is buffered to produce the LO signal and then mixed on chip with the RF signal to obtain an IF frequency at the MIXER output pins This IF frequency is typically chosen to be 10 7MHz although placement at 11 50MHz can further enhance AM mode performance and minimize IF circuitryThe PLL provides phase comparator reference frequencies of 10 12 5 25 and 100kHz The tuning resolutions result-ing from these reference frequencies are determined by di-viding the reference by the premix modulus Table II shows the tuning resolutions possibleThe DS8911contains the following logic elements a volt-age controlled oscillator a reference oscillator a 14-bit pro-grammable dual-modulus counter a reference frequency di-vider chain a premix divider a mixer a phase comparator a charge pump an operational amplifier and control circuitry for latched serial data entryThe DS8913includes all the above logic elements except that it requires a 10MHz reference frequency instead of 12MHzFeaturesY Direct synthesis of LW MW SW FM and WB frequenciesY Serial data entry for simplified processor control Y 10 12 5 25 and 100kHz reference frequencies Y 8possible tuning resolutions (see Table II)Y An op amp with high impedance inputs for loop filtering YProgrammable mixer with high dynamic rangeTRI-STATE is a registered trademark of National Semiconductor Corporation MICROWIRE TM is a trademark of National Semiconductor Corporation C 1995National Semiconductor CorporationRRD-B30M105 Printed in U S AConnection DiagramPlastic Chip CarrierTL F 7398–8Top ViewOrder Number DS8911V DS8913VSee NS Package Number V28APin DescriptionsV CC1 The V CC1pin provides a5V supply source for all circuitry except the reference divider chain op amp and mix-er sections of the dieV CC2 The V CC2pin provides a12V supply source for the Op ampV CCL The V CCL pin provides an isolated5V supply source for the premix divider and mixer functionsV CCM The V CCM pin provides a5V supply source for the reference oscillator and divider chain down through the50 Hz output thus enabling low standby current for time-of-day clock applicationsGND1 GND2 GNDL and GNDM Provide isolated circuit ground for the various sections of the deviceDATA and CLOCK The DATA and CLOCK inputs are for serial data entry from a controller They are CMOS inputs with TTL logic thresholds The24-bit data stream is loaded into the PLL on the positive transition of the CLOCK The first14bits of the data stream select PLL divide code in binary form MSB first The15th through24th bits select the premix modulus the reference frequency the bit output status and the test operate modes as shown in Tables I through VENABLE The ENABLE input is a CMOS input with a TTL logic threshold The ENABLE input enables data when at a logic‘‘one’’and latches data on the transition to a logic ‘‘zero’’BIT Outputs The open-collector BIT outputs provide either the status of shift register bits22 23 and24or enable access to key internal circuit test nodes The mode for the bit outputs is controlled by shift register bits20and21 In operation the bit outputs are intended to drive radio func-tions such as gain mute and AM FM status These outputs can also be used to program the loop gain by connection of an external resistor to IPROG Bit24output can also be used as a300millisecond timer under control of shift regis-ter bit19 During service testing these pins can be used for the purpose of either monitoring or driving internal logic points as indicated in the TEST MODES description under Table VVCOb and VCOe The Voltage Controlled Oscillator inputs drive the14-bit programmable counter and the premix divid-er These inputs are the base and emitter leads of a transis-tor which require connection of a coil varactor and several capacitors to function as a Colpitts oscillator The VCO is designed to operate up to225MHz The VCO’s minimum operating frequency may be limited by the choice of refer-ence frequency and the961minimum modulus constraint of the31 32dual modulus counterRF a and RF b The Radio Frequency inputs are fed differ-entially into the mixerIMXR The bias current for the mixer is programmed by con-nection of an external resistor to this pin The total mixer output current equals4times the current entering this pin MIXER and MIXER The MIXER outputs are the collectors of the double balanced pair mixer transistors They are in-tended to operate at voltages greater than V CC1OSCb and OSCc The Reference Oscillator inputs are part of an on-chip Pierce oscillator designed to work in conjunc-tion with2capacitors and a crystal resonator The DS8911 requires a12MHz crystal to derive the reference frequen-cies shown in Table II The DS8913requires10MHz crystal The12MHz OSC signal is also used externally as the2nd AM LO to obtain a450kHz2nd IF frequency in the AM mode2MHz The2MHz output is provided to drive a controller’s clock input50Hz The50Hz output is provided as a time reference for radios with time-of-day clocksIPROG The IPROG pin enables the charge pump to be programmed from0 25mA to1 0mA by connection of an external resistor to groundCPO The Charge Pump Output circuit sources current if the VCO frequency is high and sinks current if the VCO frequen-cy is low The CPO is wired directly to the negative input of the loop filter op ampOP AMP The OP AMP output is provided for loop filtering The op amp has high impedance PMOS gate inputs and is wired as a transconductance amplifier filter The op amp’s positive input is internally referenced while its negative input is common with the CPO output2Reference TablesTABLE IBit15Premix Modulus0d11d10TABLE IIBit Reference Tuning Resolution 1617Frequency d1Premix d10Premix 0010kHz10kHz1kHz 0112 5kHz12 5kHz1 25kHz 1025kHz25kHz2 5kHz 11100kHz100kHz10kHzTABLE IIIBit18Mode0Normal Operation1Production Test Mode OnlyThe user should always load Bit18lowTABLE IVBit19Timer0Bit24Status1Bit24for300msTIMER OPERATIONThe timer function is provided for use as a retriggerable ‘‘one shot’’to enable muting for approximately300millisec-onds after station changes The timer is enabled at bit24’s output if the normal operating mode is selected(shift regis-ter bits20and21e‘‘LOW’’)and shift register bit19data is latched as a‘‘HI’’ The timer’s output state will invert imme-diately upon latching bit19‘‘HI’’and remain inverted for approximately300milliseconds If the user readdresses the device with bit19data‘‘LOW’’before the timer finishes itscycle the timer’s BIT24output will finish out the300ms pulse Readdressing the device with bit19‘‘HI’’before the timer finishes its cycle will extend the BIT24output pulse width by300ms Addressing should be performed immedi-ately after the50Hz output transitions‘‘HI’’ BIT24’s output state is not guaranteed during the first300ms after V CC1 power up as a result of a timer reset in progressTABLE VBit FUNCTION OF2021PINS3 4 500Status of Bits22-2401Test mode110Test mode211Test mode3 TEST MODE OPERATIONTest Mode1 Enables the BIT output pins to edge trigger the phase comparator inputs and monitor an internal lock detector BIT22negative edge triggers the reference divid-er input of the phase comparator if the reference divider state is low BIT23provides the open collector ORing of the phase comparator’s pump up and down outputs BIT24 negative edge triggers the N counter input of the phase comparator if the N counter state is preconditioned lowTest Mode2 Enables the BIT outputs to clock the pro-grammable N counter monitor its output and force either its load or count condition BIT22provides the N counter out-put which negative edge triggers the phase comparator and which appears low one N counter clock pulse before it re-loads BIT23positive edge triggers the N counter’s clock input if the prescaler’s output is preconditioned HI BIT24 clears the N counter output so that loading will occur on the next N counter clock edgeTest Mode3 Enables the BIT outputs to clock the50Hz and10kHz reference dividers and monitor the reference divider input to the phase comparator BIT22positive edge clocks the10kHz reference divider chain if the10kHz out-put is preconditioned HI BIT23positive edge clocks the50 Hz divider chain BIT24is the reference divider negative edge trigger input to the phase comparator3Absolute Maximum Ratings(Note1)If Military Aerospace specified devices are required please contact the National Semiconductor Sales Office Distributors for availability and specifications Supply VoltageV CCM7V V CC17V V CC215V Input Voltage7V Output VoltageLogic7V Op Amp and Mixer Outputs15V ESD Sensitivity1000V Storage Temperature Range b65 C to a150 C Lead Temp (Soldering 10seconds)300 C Operating ConditionsMin Max Units V CCM3 55 5VV CC14 55 5VV CC27 012 0V Temperature T A b40a85 C Mixer I BIAS(Mixer a Mixer Current)120mADC Electrical Characteristics(Notes2and3)Symbol Parameter Test Conditions Min Typ Max Units V IH Logic‘‘1’’Input Voltage2 0VV IL Logic‘‘0’’Input Voltage0 8VI IH Logic‘‘1’’Input Current V IN e5 5V10m A I I Logic‘‘1’’Input Current Data Clock and Enable Inputs V IN e7V100m A I IL Logic‘‘0’’Input Current Data Clock and Enable Inputs V IN e0V b10m A V OH Logic‘‘1’’2MHz I OH e b20m A V CCM b0 3V Output VoltageI OH e b400m A V CCM b2VOp Amp I OH e b1 0mA V CC2b1 5VV OL Logic‘‘0’’2MHz I OL e20m A0 3V Output VoltageI OL e400m A0 4V50Hz I OL e250m A0 3VBit Outputs I OL e1mA0 3VOp Amp I OL e1 0mA1 5VV BIAS Op Amp Input V D Op Amp I O Shorted V CC1e5 5V V CC2e12V 200mVCPO e TRI-STATE Op Amp I OH vs I OL AppliedI CEX High Level Bit Outputs V CC1e4 5V V O e8 8V100m AOutput Current50Hz V CCM e3 5V V O e5 5V10m AMixers V CCL e V CC1e4 5V V O e12V100m A I CPO Charge Pump Program0 25mA k I CPO k1 0mA Pump-up b302I PROG a30%Current2I PROG e V CC1 R PROGPump-down b302I PROG a30%Measured I PROG to CPOTRI-STATE0100nA I CCM V CCM Supply Current V CCM e5 5V OSCC e High0 51 0mA(Static)I CC1a V CC1a V CCL V CC e5 5V Bits Hi I MXR and I PROG Open2535mA I CCL Supply CurrentI CC2V CC2Supply Current V CC2e12V1 52 5mA Mixer Mixer a Mixer V CC1e V CCL e5 5V Mixer e Mixer e12V b254IMXR a25% I BIAS Current(Note4)RF IN Mixer Input Max Mixer I BIAS e20mA300mVrms Signal Level RF a or RF b Signal LevelNote1 ‘‘Absolute Maximum Ratings’’are those values beyond which the safety of the device cannot be guaranteed Except for‘‘Operating Temperature Range’’ they are not meant to imply that the device should be operated at these limitsNote2 Unless otherwise specified min max limits apply across the b40 C to a85 C temperature rangeNote3 All currents into device pins are shown as positive out of device pins as negative all voltage referenced to ground unless otherwise noted All values shown as maximum or minimum on absolute value basisNote4 Total mixer output current(Mixer a Mixer)j4times the current into the I MXR pin4AC Electrical Characteristics(Note2)Symbol Parameter Conditions Min Typ Max Units t r20%–80%Rise Time200ns t f80%–20%Fall Time200ns DATA SU Data Setup Time100ns DATA H Data Hold Time100nsV CC1e4 5V to5 5VEN SU Enable Setup Time100ns EN H Enable Hold Time100ns EN PW a Enable Positive Pulse Width200ns CLK PW a Clock Positive Pulse Width200ns CLK PW b Clock Negative Pulse Width200ns VCO f max VCO Max Frequency See Typical Wiring Diagram20225MHz OSC f max Reference Oscillator V CCM e3 5V12MHz Max FrequencyTiming DiagramTL F 7398–10 MICROWIRE TM Bus FormatTL F 7398–195TABLE VI DS8911Tuning CharacteristicsIF Tuning VCOPremix Reference TuningImageMode Frequency Range RangeModulus Frequency Resolution(MHz)(MHz)(MHz)(MHz)(kHz)(kHz)LW11 55 450 145– 290112 4–114 11010122–23 MW11 55 450 515–1 6199 4–110 21010 12 5 25 1001 1 25 2 5 1021–23 SW11 55 4505 94–6 253 5to56 11010 12 5 251 1 25 2 528–30 FM10 787 4–108 198 1–118 8110 12 5 25 10010 12 5 25 100109–130 WB10 7162 4–162 6151–152112 5 2512 5 25140–142 TV110 759 75–87 7570 45–98 451252581–109 TV210 7179 75–215 75169 1–205 112525158–194Input and Output SchematicsTL F 7398–11TL F 7398–126Input and Output Schematics (Continued)TL F 7398–13TL F 7398–14TL F 7398–15TL F 7398–16TL F 7398–17TL F 7398–187Logic DiagramDS8911 DS8913PLL SynthesizerTL F 7398–4 Note1 The14bit programmable N counter is a dual modulus counter with31 32prescaler The minimum continuous modulus of the N counter is 961 (There are a limited number of valid modulus codes below961 )The DS8913has d5Typical Application DiagramAM FM ETR Radio ApplicationTL F 7398–58Wiring DiagramsConfiguration Using PLL and First Mixer FunctionsTL F 7398–20Configuration Using PLL with LO Bypassing MixerThe mixer is de-biased by the100X resistor on the RF apin to act as an output bufferfor the LO signalTL F 7398–219D S 8911 D S 8913A M F M T V S o u n d U p -C o n v e r s i o n F r e q u e n c y S y n t h e s i z e rPhysical Dimensions inches (millimeters)Lit 103661Plastic Chip Carrier (V)Order Number DS8911V DS8913VNS Package Number V28ALIFE SUPPORT POLICYNATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF NATIONAL SEMICONDUCTOR CORPORATION As used herein 1 Life support devices or systems are devices or 2 A critical component is any component of a life systems which (a)are intended for surgical implant support device or system whose failure to perform can into the body or (b)support or sustain life and whose be reasonably expected to cause the failure of the life failure to perform when properly used in accordance support device or system or to affect its safety or with instructions for use provided in the labeling can effectivenessbe reasonably expected to result in a significant injury to the userNational Semiconductor National Semiconductor National Semiconductor National Semiconductor CorporationEuropeHong Kong LtdJapan Ltd1111West Bardin RoadFax (a 49)0-180-530858613th Floor Straight Block Tel 81-043-299-2309。

Sensors and Actuators A:Physical1.IntroductionAt present several wireless capsule endoscopy systems areavailable on the market(Given Imaging,Olympus EndoCapsule)[1].Although appealing to the patient for comfort reasons,they lack three major properties:adequate image resolution (256×256 pixels),sufficient frame rate(2–7 frames per second (fps)),and the ability to move around in a controlled way through the GI tract.These shortcomings hamper their breakthrough with gastro-enterologists,who still prefer the traditional endoscopes[2].These limitations are a direct consequence of the finite energy supply available in these capsular endoscopes.All of them being battery powered,their lifetime is limited between 6 and 8 h,consuming 25 mW.The tight energy limit does not exist in the case of inductive powering[3,25].Without the energy constraint,a higher resolution sensor and higher frame rates become possible.However,in order to get the mass of image data outside the body to the receiver,a high speed data transmitter is required.2.Requirements2.1.Data rateModerm wired endoscopes are already equipped with High-Definition(HD)CCD cameras, providing up to 30 fps at 1920×1080 pixels per frame[4].This resolution would requirea raw Bayer data rate of 78 MByte/D image sensors are not suitable for use in capsular endoscopes because of their high power consumption compared to their CMOS equivalent.HD resolution in wireless endoscopy,even highly compressed,sounds like a fairy tale,for the simple reason that high data rate and low power is hard to combine.It is a questionable prognosis if the big advantage of patient comfort will surpass the need for HD image resolution.Compared to the presently used 256×256 pixel resolution,a big improvement in image quality can already be obtained by using a 640×480 pixel(VGA)image sensor.For 10 fps,which is a major step ahead of the current 2 fps,a raw Bayer data rate of 3.84 MByte/s is required.Still being too high for low power transmission,appropriate compression algorithms have to be used to reduce the raw data rates to acceptable levels suitable for low power transmission.pressionImage compression basically removes visually redundant information from a picture or video stream,without exaggerated loss of detail or introduction of compression artifacts. The (lossy) compression algorithms are either based on removal of high frequency image content(e.g.JPEG)or on removing redundancy from the image colors[5].A20-fold compression can easily be reached without disturbing artifacts or visual image degradation.This will relax the data rate requirement to a more feasible 1.5 Mbps.The choice of the compression engine is important at system level design:depending on the type of compression more bulky and power consuming RAM is needed.2.3.Carrier frequency and modulationLittle research has been done on the choice of the carrier frequency for through-body wireless near-field transmission.The paper of Johnson and Guy describes the attenuation of electromagnetic(EM)waves through the human body[6].This paper suggests to choose a relatively low(<200 MHz)carrier,as the attenuation increases exponentially with the carrier frequency.The reduced attenuation would lead to less required transmitted power.It is currently not known if the same behavior can be expected for near fields,so it is hard to draw a definitive conclusion on the choice of frequency.The work of Chirwa et al.suggests using a carrier between 450 and 900 MHz for maximum radiation[7].This work is based on finite-difference time-domain(FDTD)simulations on a human body model,and does not include antenna loading effects.The same antenna model was used for all frequencies,which partly accounts for the lower radiation intensities at lower frequencies.This work confirms the rapidly increasing absorption above 500 MHz,both for near-and far-fields,as described by Johnson[6].The experiments of Wang et al.describe the in-vitro characterization of ingestible capsules for 30 MHz and 868 MHz[8].They conclude that the low frequency capsule is less influenced by surrounding tissues,shows less orientation-dependent fading and a higher signal to noise(S/N)ratio for a certain power consumption.The research suggests to use a carrier below 500 MHz,although there must exist a trade off between antenna size and wavelength.For far fields,a higher carrier frequency would require a smaller antenna,so less occupied space inside the capsule.As we are working in,or close to the near-field region(the transmission and reception antenna distance is smaller or equal to the wavelength),it is not sure whether this trade-off is still valid.Near-field simulations and experiments will lead to an optimal choice of the carrier.Governmental(FCC and ERO)regulations will eventually define which frequency band can be used[9,10].Candidates could be ISM(433.05–434.79 MHz)or MICS(402–405 MHz),although the defined bandwidth limits the maximum data rate.Currently the 2 m amateur band(144 MHz)is used for data transmission,as this limits the interference with other important bands.FSK is chosen as modulation type,for its simplicity in modulation and demodulation,and its inherent insensitivity to systemnon-linearities.2.4.AntennasThe space around the transmitter antenna can be divided into two main regions as illustrated in Fig.1:far field and near field.In the far field,electric and magnetic fields propagate outward as an electromagnetic wave and are perpendicular to each other and to the direction of propagation.The angular field distribution does not depend on r,the distance from the antenna.The fields are uniquely related to each other viafree-space impedance and decay as 1/r.An the near field,the field components have different angular and radial dependence(e.g.1/r3).The near-field region includes two sub-regions:radiating,where the angular field distribution is dependent on the distance,and reactive,where the energy is stored but not radiated.For antennas whose size is comparable to wavelength(as used in UHF RFID),the approximate boundary between the far-field and the near-field region is commonly given as r=2*D2/λ,where D is the maximum antenna dimension and λis the wavelength.For electrically small antennas(as used in LF/HF RFID and in this application),the radiating near-field region is small and the boundary between the far-field and the near-field regions is commonly given as r=/2π.When the receiver antenna is located in the near field of the transmitter antenna,the coupling between the antennas affects the impedance of both antennas as well as the field distribution around them.The equivalent antenna performance parameters (gain and impedance)can no longer be specified independently from each other and become position and orientation-dependent [11].In general,to calculate the power received by the receiver in such a situation,one needs to perform a three-dimensional electromagnetic simulation of the near-field problem except when the transmitter is small and does not perturb the field of the receiving antenna.The near field of a transmitter antenna can have several tangential and radial electric and magnetic field components which can all contribute to coupling.Two ultimate cases are magnetic (inductive) coupling and electric (capacitive) coupling.In magnetic RFID systems,both receiver and transmitter antennas are coils,inductively coupled to each other like in a transformer.If the transmitter antenna is small,the coupling coefficient is proportional to[12]:where f is the frequency,N is the number of turns in transmitter coil,S is the cross-section area of the coil,B is magnetic field at the transmitter and?is the coil misalignment loss.The primary coupling mechanism in near-field transmission can be either magnetic(inductive)or electric(capacitive).Depending on the environment,the field distribution can be affected by the presence of various objects.Inductive coupled systems,where most reactive energy is stored in the magnetic field,are mostly affected by objects with high magnetic permeability.The magnetic permeability of biological tissue is practically equal to the magnetic permeability of air.On the other hand,capacitive coupling systems,where most reactive energy is stored in electric field,are affected by objects with high dielectric permittivity and loss.Since the body has high epsilon,it seems that inductive coupling is much more efficient for this kind of applications.3.Design strategy3.1.TransmitterThe first work on endoscopic telemetric capsules dates from 1957,where Mackay used a single transistor Hartley oscillator as transmitter[13].The capsule was used to measure gastric pressures,where a pressure sensitive membrane connected to an iron core modulated the oscillator inductance and thereby the oscillation frequency.The operating frequency of this device was in the 100 kHz range.The transmitter developed in earlier work[14]was redesigned such that the power consumption dropped with 66%while doubling the data rate.The transmitter consists of a single transistor Colpitts oscillator as shown in Fig.2.The circuit is built up as a common collector circuit,with the antenna connected to the collector side.The presented topology operates the transistor with a unique double function:(1)it provides gain to the feedback loop to sustain the oscillation and(2)it provides acascode function at the collector.During the capsule GI tract transit,the parasitic loads seen by the antenna change continuously.The cascode limits the antenna load detuning of the LC tank,which greatly improves frequency stability,and therefore reception.FSK modulation is achieved through modulation of the transistor base current.A change in the base current causes a change in base-emitter voltage,which influences the depletion layer width of the base-emitter junction.In this way the base-emitter capacitance is modulated,controlling the tuning of the LC tank.The maximal data rate of this transmitter is limited by the RC time constant of the R data resistor and the capacitance seen at the base.It is clear that from a frequency higher than 1/(R data*C base),the modulation index decreases,because the injected base current is shorted in the base capacitance.Although the occupied bandwidth decreases,the S/N ratio decreases too,and robust demodulation becomes more difficult at faster modulation rates.From experiments,the limit was found to be at 2 Mbps.As stated earlier in the requirements,it is more beneficial to use inductive near-field coupling for the transmitter–receiver system. For that purpose a coil transmitter antenna was developed,which was tuned to resonance around the transmitter frequency,using a trimmable capacitor.The tuned regime(for maximum power output)was measured using a spectrum analyzer.The antenna was designed as a proof of concept,but not simulated or optimized.For that reason further optimization regarding coupling,coil layout and matching is required in the future,when dedicated antennas will be designed and simulated.3.2.ReceiverThe receiver of[14]was reused,which is based on a SA639 IC application note.It consists of an IF mixing system used in DECT receivers.The FSK modulated carrier is converted back to a serial data stream by mixing the IF carrier with a phase-shifted version of itself.This results in a DC component containing the data and a component at twice the IF frequency,which is attenuated using a low pass filter.A2 m low noise amplifier(LNA)was built and included in the system,to improve the noise figure(NF)of the receiving system.The SA639 receiver has a NF of 11 dB,where the LNA has NF of3 dB with a gain of 20 dB.By including the LNA,the overall NF decreased with 6 dB.A tuned loop was used as receiving antenna,designed as a proof of concept for near-field reception.The received data is resynchronized in a hardware developed data recovery system,written in VHDL and implemented in a Xilinx Spartan XC3S200.When a synchronization pattern for a newimage frame is detected in the data stream,the data is buffered in a FIFO and sent through USB to a PC.On the PC the image data is visualized on the screen.The receiver setup is depicted in Fig.3,showing from left to right the USB interface+data recovery,the FSK demodulation board with the PLL below,and the LNA+loop receiver antenna.4.SimulationsThe transmitter circuit was simulated using ELDO[23],in order to:•optimize the component values to maximize the output power,•optimize the component values for the correct carrier frequency,•check for stable and fast start-up behavior,•check the influence of parasitic(capacitive)load changes on the output,•optimize the frequency deviation depending on the DA TA input.The sweep simulation depicted in Fig.4 shows the output frequency at DC modulation.The typical frequency deviation between DA TA=0 and DA TA=1 is approximately 180 kHz.The simulation results in Fig.5 shows the modulated frequency of the transmitter with the DA TA input toggling at 2 Mbps.The short nanosecond spikes are a result of the frequency measurement algorithm.Note the exponential curves at each data edge.This is the consequence of the base capacitance being(dis)charged through the R data resistor.This RC time constant puts a limit on the maximum achievable data rate.An important advantage of this transmitter topology is the insensitivity of the frequency vs.load capacitance variations(Fig.6).The change in frequency is<0.15%for a change in capacitive load of 20 pF.This is a huge improvement compared to[13],where the output frequency is directly proportional to the output capacitive load.5.MeasurementsThe transmitter–receiver system functionality was initially characterized using the setup as described in[24].This setup showed an unmodulated output power of -18 dBm at 50,for a supply voltage of 1.8 V.The power consumption is 2 mW at a modulation rate of 2 Mbps,being equivalent to 15–20 VGA frames/s,using appropriate compression.This results in a FOM of 1 nJ/bit,which is lower than most state-of-the-art transmitter circuits,see Table 1.Fig.7 depicts the data output of the receiver together with the data from the pseudorandom generator,showing that we are able to transmit and receive a data stream a 2 Mbps.A more qualitative test setup was conceived in the meantime, depicted in Figs.8 and 9.The transmitted data is generated from a Micron MT9V013 VGA image sensor(Fig.8 top board+sensor were provided by V ector partner Neuricam).As 30 VGA frames per second are continuously provided,and maximally 2 Mbps can be handled by the transmitter,only one out of 30 frames is buffered in a DRAM,and then slowly released from the buffer(Fig.8,bottom board).This leads to a data rate of 1.5 Mbps,or about 0.5 VGA fps which is fed to the transmitter.To avoid ground loops,or influence/crosstalk between the transmitter and the data generation board,a galvanic isolation between both is mandatory.A TOSLINK optical transmitter and receiver(TOTX147 and TORX147)were used for this purpose,allowing complete galvanic isolation between the data source and the transmitter.Fig.9 depicts the functional transmitter/receiver setup,together with the GUI showing the received picture.Transmitter and receiver are at a distance of about 20 cm.The big difference between[24] and this setup is the use of real VGA image data,as well as the use of near field instead of far-field antennas.For these tests,the transmitter was battery powered by a 3 V coin cell.The transmitter measures only 0.42 cm2 which easily enablesintegration into an endoscopic capsule.Fig.10 depicts the assembled FSK transmitter.6.ConclusionA simple,high data rate and low power transmitter is designed,fabricated and measured.It enables a 4–5 times higher image resolution(VGA instead of QVGA)and 15–17 fps for inductively powered endoscopic capsules.This development greatly improves the acceptation level by the surgeon,and will help to achieve correct diagnosis for GI tract diseases.Future improvements and measurements consist of improving the Tx-Rx antenna system,full integration with a wireless powering module and quantitative tests of the transmitter module,like Bit Error Rate(BER)vs.distance and BER vs.transmission power.More tests need be done on the functional integration of an inductive powering module and transmitter,especially with regards to harmonics of the inductive power link disturbing the transmitted FSK spectrum.AcknowledgmentsThis work is supported by the European Community,within the 6th Framework Programme, through the V ector project(contract number 0339970).The authors wish to thank the project partners and the funding organization.。

RECOMMENDED OPERATING CONDITIONS
Rating Power Supply Voltage Maximum 1st IF Maximum 2nd IF Ambient Temperature Range Pin 4, 19 – – – Symbol VCC fIF1 fIF2 TA Value 2.0 to 6.0 21 3.0 – 40 to + 85 Unit Vdc MHz MHz °C
15 Op Amp In – 14 Limiter Output 13 Quad Input
Each device contains 142 active transistors.
© Motorola, Inc. 1996
Rev 3
MOTOROLA ANALOG IC DEVICE DATA
1
MC13135 MC13136
24 1
DW SUFFIX PLASTIC PACKAGE CASE 751E (SO–24L)
ORDERING INFORMATION
Device MC13135P MC13135DW MC13136P MC13136DW TA = – 40° to +85°C Operating Temperature Range Package Plastic DIP SO–24L Plastic DIP SO–24L
7
19 VCC2 18 2nd Mixer In
AF
8 9 17 16
AF
8 9 17 Audio Out 16 Buffered RSSI Output
Demod Decouple 1 10 Decouple 2 11 RSSI 12 Limiter

More informationPhase Noise and Frequency Stability in OscillatorsPresenting a comprehensive account of oscillator phase noise and frequency stability,this practical text is both mathematically rigorous and accessible.An in-depth treatmentof the noise mechanism is given,describing the oscillator as a physical system,andshowing that simple general laws govern the stability of a large variety of oscillatorsdiffering in technology and frequency range.Inevitably,special attention is given to am-plifiers,resonators,delay lines,feedback,andflicker(1/f)noise.The reverse engineeringof oscillators based on phase-noise spectra is also covered,and end-of-chapter exercisesare given.Uniquely,numerous practical examples are presented,including case studiestaken from laboratory prototypes and commercial oscillators,which allow the oscillatorinternal design to be understood by analyzing its phase-noise spectrum.Based on tuto-rials given by the author at the Jet Propulsion Laboratory,international IEEE meetings,and in industry,this is a useful reference for academic researchers,industry practitioners,and graduate students in RF engineering and communications engineering.Additional materials are available via /rubiola.Enrico Rubiola is a Senior Scientist at the CNRS FEMTO-ST Institute and a Professorat the Universit´e de Franche Comt´e.With previous positions as a Professor at theUniversit´e Henri Poincar´e,Nancy,and in Italy at the University Parma and thePolitecnico di Torino,he has also consulted at the NASA/Caltech Jet PropulsionLaboratory.His research interests include low-noise oscillators,phase/frequency-noisemetrology,frequency synthesis,atomic frequency standards,radio-navigation systems,precision electronics from dc to microwaves,optics and gravitation.More informationThe Cambridge RF and Microwave Engineering SeriesSeries EditorSteve C.CrippsPeter Aaen,Jaime Pl´a and John Wood,Modeling and Characterization of RF andMicrowave Power FETsEnrico Rubiola,Phase Noise and Frequency Stability in OscillatorsDominique Schreurs,M´a irt´ın O’Droma,Anthony A.Goacher and Michael Gadringer,RF Amplifier Behavioral ModelingFan Y ang and Y ahya Rahmat-Samii,Electromagnetic Band Gap Structures in AntennaEngineeringForthcoming:Sorin V oinigescu and Timothy Dickson,High-Frequency Integrated CircuitsDebabani Choudhury,Millimeter W aves for Commercial ApplicationsJ.Stephenson Kenney,RF Power Amplifier Design and LinearizationDavid B.Leeson,Microwave Systems and EngineeringStepan Lucyszyn,Advanced RF MEMSEarl McCune,Practical Digital Wireless Communications SignalsAllen Podell and Sudipto Chakraborty,Practical Radio Design TechniquesPatrick Roblin,Nonlinear RF Circuits and the Large-Signal Network AnalyzerDominique Schreurs,Microwave Techniques for MicroelectronicsJohn L.B.Walker,Handbook of RF and Microwave Solid-State Power AmplifiersPhase Noise and Frequency Stability in OscillatorsENRICO RUBIOLAProfessor of Electronics FEMTO-ST Institute CNRS and Universit´e de Franche Comt´e Besanc ¸on,FranceMore informationMore informationCAMBRIDGE UNIVERSITY PRESSCambridge,New Y ork,Melbourne,Madrid,Cape Town,Singapore,S˜a o Paulo,DelhiCambridge University PressThe Edinburgh Building,Cambridge CB28RU,UKPublished in the United States of America by Cambridge University Press,New Y orkInformation on this title:/9780521886772C Cambridge University Press2009This publication is in copyright.Subject to statutory exceptionand to the provisions of relevant collective licensing agreements,no reproduction of any part may take place withoutthe written permission of Cambridge University Press.First published2009Printed in the United Kingdom at the University Press,CambridgeA catalog record for this publication is available from the British LibraryISBN978-0-521-88677-2hardbackCambridge University Press has no responsibility for the persistence oraccuracy of URLs for external or third-party internet websites referred toin this publication,and does not guarantee that any content on suchwebsites is,or will remain,accurate or appropriate.More informationContentsForeword by Lute Maleki page ixForeword by David Leeson xiiPreface xv How to use this book xviSupplementary material xviii Notation xix 1Phase noise and frequency stability11.1Narrow-band signals11.2Physical quantities of interest51.3Elements of statistics91.4The measurement of power spectra131.5Linear and time-invariant(LTI)systems191.6Close-in noise spectrum221.7Time-domain variances251.8Relationship between spectra and variances291.9Experimental techniques30Exercises33 2Phase noise in semiconductors and amplifiers352.1Fundamental noise phenomena352.2Noise temperature and noisefigure372.3Phase noise and amplitude noise422.4Phase noise in cascaded amplifiers492.5 Low-flicker amplifiers522.6 Detection of microwave-modulated light62Exercises65 3Heuristic approach to the Leeson effect673.1Oscillator fundamentals673.2The Leeson formula72More informationvi Contents3.3The phase-noise spectrum of real oscillators753.4Other types of oscillator824Phase noise and feedback theory884.1Resonator differential equation884.2Resonator Laplace transform924.3The oscillator964.4Resonator in phase space1014.5Proof of the Leeson formula1114.6Frequency-fluctuation spectrum and Allan variance1164.7 A different,more general,derivation of the resonatorphase response1174.8 Frequency transformations1215Noise in delay-line oscillators and lasers1255.1Basic delay-line oscillator1255.2Optical resonators1285.3Mode selection1305.4The use of a resonator as a selectionfilter1335.5Phase-noise response1385.6Phase noise in lasers1435.7Close-in noise spectra and Allan variance1455.8Examples1466Oscillator hacking1506.1General guidelines1506.2About the examples of phase-noise spectra1546.3Understanding the quartz oscillator1546.4Quartz oscillators156Oscilloquartz OCXO8600(5MHz AT-cut BV A)156Oscilloquartz OCXO8607(5MHz SC-cut BV A)159RAKON PHARAO5MHz quartz oscillator162FEMTO-ST LD-cut quartz oscillator(10MHz)164Agilent10811quartz(10MHz)166Agilent noise-degeneration oscillator(10MHz)167Wenzel501-04623(100MHz SC-cut quartz)1716.5The origin of instability in quartz oscillators1726.6Microwave oscillators175Miteq DRO mod.D-210B175Poseidon DRO-10.4-FR(10.4GHz)177Poseidon Shoebox(10GHz sapphire resonator)179UWA liquid-N whispering-gallery9GHz oscillator182More informationContents vii6.7Optoelectronic oscillators185NIST10GHz opto-electronic oscillator(OEO)185OEwaves Tidalwave(10GHz OEO)188 Exercises190Appendix A Laplace transforms192References196Index202More informationForeword by Lute MalekiGiven the ubiquity of periodic phenomena in nature,it is not surprising that oscillatorsplay such a fundamental role in sciences and technology.In physics,oscillators are thebasis for the understanding of a wide range of concepts spanningfield theory and linearand nonlinear dynamics.In technology,oscillators are the source of operation in everycommunications system,in sensors and in radar,to name a few.As man’s study ofnature’s laws and human-made phenomena expands,oscillators have found applicationsin new realms.Oscillators and their interaction with each other,usually as phase locking,and withthe environment,as manifested by a change in their operational parameters,form thebasis of our understanding of a myriad phenomena in biology,chemistry,and evensociology and climatology.It is very difficult to account for every application in whichthe oscillator plays a role,either as an element that supports understanding or insight oran entity that allows a given application.In all thesefields,what is important is to understand how the physical parametersof the oscillator,i.e.its phase,frequency,and amplitude,are affected,either by theproperties of its internal components or by interaction with the environment in whichthe oscillator resides.The study of oscillator noise is fundamental to understanding allphenomena in which the oscillator model is used in optimization of the performance ofsystems requiring an oscillator.Simply stated,noise is the unwanted part of the oscillator signal and is unavoidablein practical systems.Beyond the influence of the environment,and the non-ideality ofthe physical elements that comprise the oscillator,the fundamental quantum nature ofelectrons and photons sets the limit to what may be achieved in the spectral purity of thegenerated signal.This sets the fundamental limit to the best performance that a practicaloscillator can produce,and it is remarkable that advanced oscillators can reach it.The practitioners who strive to advance thefield of oscillators in time-and-frequencyapplications cannot be content with knowledge of physics alone or engineering alone.The reason is that oscillators and clocks,whether of the common variety or the advancedtype,are complex“systems”that interact with their environment,sometimes in waysthat are not readily obvious or that are highly nonlinear.Thus the physicist is needed toidentify the underlying phenomenon and the parameters affecting performance,and theengineer is needed to devise the most effective and practical approach to deal with them.The present monograph by Professor Enrico Rubiola is unique in the extent to which itsatisfies both the physicist and the engineer.It also serves the need to understand bothMore informationx Forewordsthe fundamentals and the practice of phase-noise metrology,a required tool in dealingwith noise in oscillators.Rubiola’s approach to the treatment of noise in this book is based on the input–output transfer functions.While other approaches lead to some of the same results,this treatment allows the introduction of a mathematical rigor that is easily tractable byanyone with an introductory knowledge of Fourier and Laplace transforms.In particular,Rubiola uses this approach to obtain a derivation,fromfirst principles,of the Leesonformula.This formula has been used in the engineering literature for the noise analysisof the RF oscillator since its introduction by Leeson in1966.Leeson evidently arrivedat it without realizing that it was known earlier in the physics literature in a differentform as the Schawlow–Townes linewidth for the laser oscillator.While a number ofother approaches based on linear and nonlinear models exist for analyzing noise inan oscillator,the Leeson formula remains particularly useful for modeling the noisein high-performance oscillators.Given its relation to the Schawlow–Townes formula,it is not surprising that the Leeson model is so useful for analyzing the noise in theoptoelectronic oscillator,a newcomer to the realm of high-performance microwave andmillimeter-wave oscillators,which are also treated in this book.Starting in the Spring of2004,Professor Rubiola began a series of limited-timetenures in the Quantum Sciences and Technologies group at the Jet Propulsion Labo-ratory.Evidently,this can be regarded as the time when the initial seed for this bookwas conceived.During these visits,Rubiola was to help architect a system for themeasurement of the noise of a high-performance microwave oscillator,with the sameexperimental care that he had previously applied and published for the RF oscillators.Characteristically,Rubiola had to know all the details about the oscillator,its principleof operation,and the sources of noise in its every component.It was only then that hecould implement the improvement needed on the existing measurement system,whichwas based on the use of a longfiber delay in a homodyne setup.Since Rubiola is an avid admirer of the Leeson model,he was interested in applyingit to the optoelectronic oscillator,as well.In doing so,he developed both an approachfor analyzing the performance of a delay-line oscillator and a scheme based on Laplacetransforms to derive the Leeson formula,advancing the original,heuristic,approach.These two treatments,together with the range of other topics covered,should makethis unique book extremely useful and attractive to both the novice and experiencedpractitioners of thefield.It is delightful to see that in writing the monograph,Enrico Rubiola has so openlybared his professional persona.He pursues the subject with a blatant passion,andhe is characteristically not satisfied with“dumbing down,”a concept at odds withmathematical rigor.Instead,he provides visuals,charts,and tables to make his treatmentaccessible.He also shows his commensurate tendencies as an engineer by providingnumerical examples and details of the principles behind instruments used for noisemetrology.He balances this with the physicist in him that looks behind the obvious forthe fundamental causation.All this is enhanced with his mathematical skill,of which healways insists,with characteristic modesty,he wished to have more.Other ingredients,missing in the book,that define Enrico Rubiola are his knowledge of ancient languagesMore informationForewords xi and history.But these could not inform further such a comprehensive and extremelyuseful book on the subject of oscillator noise.Lute MalekiNASA/Caltech Jet Propulsion Laboratoryand OEwaves,Inc.,February2008More informationForeword by David LeesonPermit me to place Enrico Rubiola’s excellent book Phase Noise and Frequency Stabilityin Oscillators in context with the history of the subject over the pastfive decades,goingback to the beginnings of my own professional interest in oscillator frequency stability.Oscillator instabilities are a fundamental concern for systems tasked with keeping anddistributing precision time or frequency.Also,oscillator phase noise limits the demod-ulated signal-to-noise ratio in communication systems that rely on phase modulation,such as microwave relay systems,including satellite and deep-space parablyimportant are the dynamic range limits in multisignal systems resulting from the mask-ing of small signals of interest by oscillator phase noise on adjacent large signals.Forexample,Doppler radar targets are masked by ground clutter noise.These infrastructure systems have been well served by what might now be termedthe classical theory and measurement of oscillator noise,of which this volume is acomprehensive and up-to-date tutorial.Rubiola also exposes a number of significantconcepts that have escaped prior widespread notice.My early interest in oscillator noise came as solid-state signal sources began to beapplied to the radars that had been under development since the days of the MIT RadiationLaboratory.I was initiated into the phase-noise requirements of airborne Doppler radarand the underlying arts of crystal oscillators,power amplifiers,and nonlinear-reactancefrequency multipliers.In1964an IEEE committee was formed to prepare a standard on frequency stability.Thanks to a supportive mentor,W.K.Saunders,I became a member of that group,whichincluded leaders such as J.A.Barnes and L.S.Cutler.It was noted that the independentuse of frequency-domain and time-domain definitions stood in the way of the develop-ment of a common standard.To promote focused interchange the group sponsored theNovember1964NASA/IEEE Conference on Short Term Frequency Stability and editedthe February1966Special Issue on Frequency Stability of the Proceedings of the IEEE.The context of that time included the appreciation that self-limiting oscillators andmany systems(FM receivers with limiters,for example)are nonlinear in that theylimit amplitude variations(AM noise);hence the focus on phase noise.The modestfrequency limits of semiconductor devices of that period dictated the common usage ofnonlinear-reactance frequency multipliers,which multiply phase noise to the point whereit dominates the output noise spectrum.These typical circuit conditions were secondnature then to the“short-term stability community”but might not come so readily tomind today.More informationForewords xiii Thefirst step of the program to craft a standard that would define frequency stabilitywas to understand and meld the frequency-and time-domain descriptions of phaseinstability to a degree that was predictive and permitted analysis and optimization.Bythe time the subcommittee edited the Proc.IEEE special issue,the wide exchange ofviewpoints and concepts made it possible to synthesize concise summaries of the workin both domains,of which my own model was one.The committee published its“Characterization of frequency stability”in IEEE Trans.Instrum.Meas.,May1971.This led to the IEEE1139Standards that have served thecommunity well,with advances and revisions continuing since their initial publication.Rubiola’s book,based on his extensive seminar notes,is a capstone tutorial on thetheoretical basis and experimental measurements of oscillators for which phase noiseand frequency stability are primary issues.In hisfirst chapter Rubiola introduces the reader to the fundamental statistical de-scriptions of oscillator instabilities and discusses their role in the standards.Then in thesecond chapter he provides an exposition of the sources of noise in devices and circuits.In an instructive analysis of cascaded stages,he shows that,for modulative or parametricflicker noise,the effect of cascaded stages is cumulative without regard to stage gain.This is in contrast with the well-known treatment of additive noise using the Friisformula to calculate an equivalent input noise power representing noise that may originateanywhere in a cascade of real amplifiers.This example highlights the concept that“themodel is not the actual thing.”He also describes concepts for the reduction offlickernoise in amplifier stages.In his third chapter Rubiola then combines the elements of thefirst two chapters toderive models and techniques useful in characterizing phase noise arising in resonatorfeedback oscillators,adding mathematical formalism to these in the fourth chapter.Inthefifth chapter he extends the reader’s view to the case of delay-line oscillators suchas lasers.In his sixth chapter,Rubiola offers guidance for the instructive“hacking”ofexisting oscillators,using their external phase spectra and other measurables to estimatetheir internal configuration.He details cases in which resonatorfluctuations mask circuitnoise,showing that separately quantifying resonator noise can be fruitful and that devicenoisefigure and resonator Q are not merely arbitraryfitting factors.It’s interesting to consider what lies ahead in thisfield.The successes of today’sconsumer wireless products,cellular telephony,WiFi,satellite TV,and GPS,arise directlyfrom the economies of scale of highly integrated circuits.But at the same time thisintroduces compromises for active-device noise and resonator quality.A measure ofthe market penetration of multi-signal consumer systems such as cellular telephonyand WiFi is that they attract enough users to become interference-limited,often fromsubscribers much nearer than a distant base station.Hence low phase noise remainsessential to preclude an unacceptable decrease of dynamic range,but it must now beachieved within narrower bounds on the available circuit elements.A search for new understanding and techniques has been spurred by this requirementfor low phase noise in oscillators and synthesizers whose primary character is integrationand its accompanying minimal cost.This body of knowledge is advancing througha speculative and developmental phase.Today,numerical nonlinear circuit analysisMore informationxiv Forewordssupports additional design variables,such as the timing of the current pulse in nonlinearoscillators,that have become feasible because of the improved capabilities of bothsemiconductor devices and computers.Thefield is alive and well,with emerging players eager tofind a role on the stage fortheir own scenarios.Professionals and students,whether senior or new to thefield so ablydescribed by Rubiola,will benefit from his theoretical rigor,experimental viewpoint,and presentation.David B.LeesonStanford UniversityFebruary2008More informationPrefaceThe importance of oscillators in science and technology can be outlined by two mile-stones.The pendulum,discovered by Galileo Galilei in the sixteenth century,persistedas“the”time-measurement instrument(in conjunction with the Earth’s rotation period)until the piezoelectric quartz resonator.Then,it was not by chance that thefirst inte-grated circuit,built in September1958by Jack Kilby at the Bell Laboratories,was aradio-frequency oscillator.Time,and equivalently frequency,is the most precisely measured physical quantity.The wrist watch,for example,is probably the only cheap artifact whose accuracy ex-ceeds10−5,while in primary laboratories frequency attains the incredible accuracy ofa few parts in10−15.It is therefore inevitable that virtually all domains of engineeringand physics rely on time-and-frequency metrology and thus need reference oscillators.Oscillators are of major importance in a number of applications such as wireless com-munications,high-speed digital electronics,radars,and space research.An oscillator’srandomfluctuations,referred to as noise,can be decomposed into amplitude noise andphase noise.The latter,far more important,is related to the precision and accuracy oftime-and-frequency measurements,and is of course a limiting factor in applications.The main fact underlying this book is that an oscillator turns the phase noise of itsinternal parts into frequency noise.This is a necessary consequence of the Barkhausencondition for stationary oscillation,which states that the loop gain of a feedback oscillatormust be unity,with zero phase.It follows that the phase noise,which is the integral ofthe frequency noise,diverges in the long run.This phenomenon is often referred to asthe“Leeson model”after a short article published in1966by David B.Leeson[63].Onmy part,I prefer the term Leeson effect in order to emphasize that the phenomenon isfar more general than a simple model.In2001,in Seattle,Leeson received the W.G.Cady award of the IEEE International Frequency Control Symposium“for clear physicalinsight and[a]model of the effects of noise on oscillators.”In spring2004I had the opportunity to give some informal seminars on noise in oscil-lators at the NASA/Caltech Jet Propulsion Laboratory.Since then I have given lecturesand seminars on noise in industrial contexts,at IEEE symposia,and in universities andgovernment laboratories.The purpose of most of these seminars was to provide a tuto-rial,as opposed to a report on advanced science,addressed to a large-variance audiencethat included technicians,engineers,Ph.D.students,and senior scientists.Of course,capturing the attention of such a varied audience was a challenging task.The stimu-lating discussions that followed the seminars convinced me I should write a workingMore informationxvi Prefacedocument1as a preliminary step and then this book.In writing,I have made a seriouseffort to address the same broad audience.This work could not have been written without the help of many people.The gratitudeI owe to my colleagues and friends who contributed to the rise of the ideas containedin this book is disproportionate to its small size:R´e mi Brendel,Giorgio Brida,G.JohnDick,Michele Elia,Patrice F´e ron,Serge Galliou,Vincent Giordano,Charles A.(Chuck)Greenhall,Jacques Groslambert,John L.Hall,Vladimir S.(Vlad)Ilchenko,LaurentLarger,Lutfallah(Lute)Maleki,Andrey B.Matsko,Mark Oxborrow,Stefania R¨o misch,Anatoliy B.Savchenkov,Franc¸ois Vernotte,Nan Yu.Among them,I owe special thanks to the following:Lute Maleki for giving me theopportunity of spending four long periods at the NASA/Caltech Jet Propulsion Labora-tory,where I worked on noise in photonic oscillators,and for numerous discussions andsuggestions;G.John Dick,for giving invaluable ideas and suggestions during numerousand stimulating discussions;R´e mi Brendel,Mark Oxborrow,and Stefania R¨o misch fortheir personal efforts in reviewing large parts of the manuscript in meticulous detail andfor a wealth of suggestions and criticism;Vincent Giordano for supporting my effortsfor more than10years and for frequent and stimulating discussions.I wish to thank some manufacturers and their local representatives for kindness andprompt help:Jean-Pierre Aubry from Oscilloquartz;Vincent Candelier from RAKON(formerly CMAC);Art Faverio and Charif Nasrallah from Miteq;Jesse H.Searles fromPoseidon Scientific Instruments;and Mark Henderson from Oewaves.Thanks to my friend Roberto Bergonzo,for the superb picture on the front cover,entitled“The amethyst stairway.”For more information about this artist,visit the website.Finally,I wish to thank Julie Lancashire and Sabine Koch,of the Cambridge editorialstaff,for their kindness and patience during the long process of writing this book.How to use this bookLet usfirst abstract this book in one paragraph.Chapter1introduces the language ofphase noise and frequency stability.Chapter2analyzes phase noise in amplifiers,includ-ingflicker and other non-white phenomena.Chapter3explains heuristically the physicalmechanism of an oscillator and of its noise.Chapter4focuses on the mathematics thatdescribe an oscillator and its phase noise.For phase noise,the oscillator turns out to bea linear system.These concepts are extended in Chapter5to the delay-line oscillatorand to the laser,which is a special case of the latter.Finally,Chapter6analyzes indepth a number of oscillators,both laboratory prototypes and commercial products.Theanalysis of an oscillator’s phase noise discloses relevant details about the oscillator.There are other books about oscillators,though not numerous.They can be divided intothree categories:books on radio-frequency and microwave oscillators,which generallyfocus on the electronics;books about lasers,which privilege atomic physics and classical1E.Rubiola,The Leeson Effect–Phase Noise in Quasilinear Oscillators,February2005,arXiv:physics/0502143,now superseded by the present text.PrefacexviideeperreadingbasictheoreticaladvancedtheoreticallegendexperimentalistlecturerdeeperreadingFigure1Asymptotic reading paths:on the left,for someone planning lectures on oscillatornoise;on the right,for someone currently involved in practical work on oscillators.optics;books focusing on the relevant mathematical physics.The present text is uniquein that we look at the oscillator as a system consisting of more or less complex interactingblocks.Most topics are innovative,and the overlap with other books about oscillatorsor time-and-frequency metrology is surprisingly small.This may require an additionaleffort on the part of readers already familiar with the subject area.The core of this book rises from my experimentalist soul,which later became con-vinced of the importance of the mathematics.The material was originally thought anddrafted in the following(dis)order(see Fig.1):3Heuristic approach,6Oscillator hack-ing,4Feedback theory,5Delay-line oscillators.Thefinal order of subjects aims at amore understandable presentation.In seminars,I have often presented the material in the3–6–4–5order.Y et,the best reading path depends on the reader.T wo paths are suggestedin Fig.1for two“asymptotic”reader types,i.e.a lecturer and experimentalist.Whenplanning to use this book as a supplementary text for a university course,the lecturer More information。