Novel zero-voltage and zero-current-switching full bridge PWM converter using transformer auxiliary

Zero-V oltage–Zero-Current Switching in High-Output-V oltage Full-Bridge PWM Converters Using the Interwinding CapacitanceDjordje Garabandic,William G.Dunford,Senior Member,IEEE,and Mark EdmundsAbstract—A novel zero-voltage and zero-current-switching (ZVZCS)full-bridge(FB)pulsewidth-modulated(PWM) converter is proposed.The new converter uses the interwinding capacitance and a small primary-side inductor to achieve a zero-current–zero-voltage turn off and a zero-current turn on of the passive-to-active leg transistors.The turn off of the active-to-passive leg transistors is with zero voltage,and the turn on is with zero voltage and zero current across them.The ringing caused by the parasitic interwinding capacitance and by the reverse recovery of the rectifiers is reduced.The new converter is attractive for high-output-voltage applications(600–1000V), where the interwinding capacitance is sufficiently dominant. In addition,switches such as insulated gate bipolar transistors (IGBT’s)and MCT’s can be used at higher frequencies which is particularly desirable for high-power application(above2 kW).The experimental results obtained from an IGBT-based 62.5-kHz dc/dc converter with a rated output voltage of600V and a nominal power of1.2kW are presented.I.I NTRODUCTIONP ARASITIC components such as the leakage inductance of the transformer and the output capacitance of the switches are effectively used in the zero-voltage-switching(ZVS)full-bridge(FB)pulsewidth-modulated(PWM)converter to reduce the switching losses[1]–[4].This switching technique is particularly well suited for high-frequency MOSFET-based applications.Yet,ZVS alone does not address problems related to the reverse recovery of the output rectifiers,and this is critical in high-output-voltage(HOV)converters.In addition, the use of insulated gate bipolar transistors(IGBT’s)instead of MOSFET’s is not trivial.A common characteristic of HOV-FB converters is that the number of secondary turns of the power transformer is considerably larger than the number of primary turns. Unavoidably,this increases the interwinding capacitance of the secondary.The interwinding capacitance,together with the equivalent capacitance of the high-voltage rectifiers,causes the so-called reverse-recovery ringing and a high-reverse-recovery-voltage spike across the diodes.These phenomena are the major limiting factors for the maximum achievableManuscript received April8,1998;revised July14,1998.Recommended by Associate Editor,J.Thottuvelil.D.Garabandic and M.Edmunds are with Xantrex Technology,Inc., Burnaby,B.C.,V5A4V7,Canada(e-mail:gar@).W.G.Dunford is with the Department of Electrical Engineering,University of British Columbia,B.C.,V6T1Z4,Canada.Publisher Item Identifier S0885-8993(99)01835-9.output voltage,and at the same time they are the generators of much of the noise and losses.A possible dissipative clamp that limits the reverse-recovery voltage spike is presented in[2].The optimal use of that clamp is forfixed output-voltage converters,since the clamping level changes with the output voltage.A very efficient active clamp as in[3]is less cost effective since it requires an additional transistor andfloating drivers for each rectifier stage.The commutating aid circuit consisting of a small inductor and two clamping diodes improves the performance of a ZVS converter[4].It reduces the reverse-recovery ringing,extends the soft-switching region toward the light-load conditions,and enables an optimal transformer design.The main concern with the circuit proposed in[4],when used with IGBT’s,is that the ZVS is not the preferred oper-ating mode.The efficiency of an IGBT bridge is higher when operating in the zero-current-switching(ZCS)mode.A simple circuit proposed in[5]achieves this goal with a capacitor and a saturable reactor placed in series with the primary.There,the current in the primary is reset during the freewheeling period by the series capacitor and sustained at close to zero by the saturable reactor.However,the voltage that builds up across the series capacitor additionally increases the reverse voltage across the output diodes.In HOV applications,it is imperative to keep the reverse voltage of the diodes as low as possible and the circuit as in[5]is not suitable.The proposed HOV-FB-ZVZCS-PWM converter is shown in Fig.1.The basic topology is the same as that of a ZVS-FB-PWM converter.The output consists of two stacked rectifier bridges(if necessary,this number can be larger).The addi-tional small(balancing)transformers(and)balance the terminal currents of the two secondary windings,and their function will be explained later.Here,only an outline of the main concept is given.During the active-to-passive transition, the high interwinding capacity between the two secondaries resets the primary current to zero.By the time the freewheeling period begins(turn on of the active-to-passive leg transistor), the current in the primary is practically zero.Consequently, the passive-to-active leg transistor turns off with zero current. The primaryreactor delays the current buildup and ensures that the turn on of the passive-to-active leg transistor is with zero current.The proposed concept was verified on an experimental 1.2-kW62.5-kHz IGBT-based FB converter with a rated output voltage of600V and a nominal power of1.2kW.0885–8993/99$10.00©1999IEEEFig.1.The proposedHOV-ZVZCS-FB-PWM.Fig.2.The two secondaries formed by a bifilar winding.This paper gives design considerations as well as the obtained experimental results.II.P RINCIPLEOFO PERATIONThe proposed ZVZCS converter is controlled by phase shifting.Basically,the same control circuitry can be used which already exists for the FB-ZVS-PWM converter.In order to explain the operation of the proposed converter,certain general assumptions will be made.•is large enough and can be treatedas a constant current source.•The dominant interwinding capacitance–dischargesthrough,and themiddleFig.3.Idealized waveforms from the circuit in Fig.1.Fig.4.The sequence of events during an active-to-passive transition.GARABANDIC et al .:ZERO-VOLTAGE–ZERO-CURRENT SWITCHING IN PWM CONVERTERS 345pointvoltage starts toconductassumption:is the outputcurrent transferred to theprimary side.STRUCTURE 2—–is still not discharged,a negative voltageacrossforces the currentthrough to drop to zero.At ,the current through the primary has reached zero.Thetransistoris the voltageacrossand.STRUCTURE 3—–is stillcharged;the result is that thediodesis discharged by theoutputcurrentreaches zero.At ,all secondary-side diodes are forward biased and the freewheeling period beginsassumption:(3)whereis the primary-side equivalentof––(4)The slopeof ,thus reducing the effective dutycycleby ,the turn onof–for thediodesandis(5)and the pick reverse-recoverycurrentdenotes the carrierlifetime,.Fig.5.The sequence of events during a passive-to-active transition.STRUCTURE 7—–.Theand is ringing.The ringing is reduced with the balancing transformers becauseeffectivelyis charged through the (saturable)magnetizing inductance of the balancingtransformersand ,and this effect will be explained in the next section.By adding the clampingdiodes–and the reverse recovery can bereduced by:1)increasing the damping factor of theresonating.Fig.6(a)and (b)proposes snubbers that achieve these goals.The main idea in both cases is to use balancing transformers that equalizethe terminal currents of the twosecondariesand .The incremental models of Structure 7(Fig.5)are346IEEE TRANSACTIONS ON POWER ELECTRONICS,VOL.14,NO.2,MARCH1999(a)(b)Fig.6.Snubbing the C i w with balancing transformers T bal1and T bal2:(a)limiting the inrush current with R snb1and R snb2,(b)limiting the inrush current with the magnetizing inductance of T bal1and T bal2,(c)the incremental model of the circuit from Fig.6(a),and (d)the incremental model of the circuit from Fig.6(b).Fig.7.Experimental waveforms (a)without and (b)with balancing transformers.Traces:v d (200V/div),i prim (10A/div),and time =2 s/div.shown in Fig.6(c)and (d).In the case of the circuit in Fig6(a),the differenceand thatcharges and is limited (and the ringingis damped)by the saturable magnetizing inductance of thebalancingtransformersand .Fig.7shows the experimental waveforms with and without the proposed balancing transformers.A comparison of thesewaveforms shows a considerable improvement.Each balanc-ing transformer consists of214GARABANDIC et al .:ZERO-VOLTAGE–ZERO-CURRENT SWITCHING IN PWM CONVERTERS347Fig.8.The soft-switchingregion.Fig.9.Waveforms obtained from the experimental converter traces:V a (200V/div),V b (200V/div),i prim (10A/div),and time =2 s/div.2)ZVZC turn on of the active-to-passive leg transistor.3)ZVZC turn off of the passive-to-active leg transistor.4)ZC turn on of the passive-to-active leg transistor.5)The current thatchargesis limited,and the ringing is damped by the balancing transformers.In the active-to-passive leg,the turn-off losses can befurther minimized by the use of external capacitors(and )[5].In the passive-to-active leg,the turn-off losses are practically zero.The turn-on losses are low in both legs due to the ZCS.Because of that,the proposed converter is particularly welcome for IGBT-based bridges.From Fig.4it can be concluded that in order to achieve the ZVZCS by using the interwinding capacitance,the following conditions have to bemet:and to providethe ZVS.The second inequality reflects the condition under which the interwinding capacitance is capable of resetting the primary current for theZCS.Fig.10.The magnified active-to-passive transition traces:V a (200V/div),V b (200V/div),i prim (10A/div),V Ciw (200V/div),and time =1s/div.Fig.11.The magnified passive-to-active transition traces:V a (200V/div),V b (200V/div),i prim (10A/div),V Ciw (200V/div),and time =1 s/div.The soft-switching region,as a functionof ,is plotted in Fig.8.In this plot,three distinct sectors are identified.Depending in which of the sectors is the operating point,the following can be concluded.1)Sector 1:The active-to-passive leg is ZVS.The primary current is only partially reset and still flowing during the freewheeling period.Thus,a ZV transition in the passive-to-active leg is initiated.Basically,this operating mode is the same as that of a FB-ZVS-PWM [1]converter.2)Sector 2:The conditions (7)and (8)are satisfied,and the converter is in the ZVZC operating mode.3)Sector 3:In light-load conditions,the active-to-passive leg is no longer ZV switched.The effect of the primaryinductanceon the pick primary current (turn-on current spike)can be assessed from the incremental model of the Structure 7(Fig.5).The upperbound for the primary currentovershoot(worst case348IEEE TRANSACTIONS ON POWER ELECTRONICS,VOL.14,NO.2,MARCH 1999TABLE IT HE T HEORETICALANDM EASURED P ARAMETERSOF THECONVERTERFig.12.Efficiency measurements:IGBT versus MOSFET.no damping)is(9)Combining (6),(8),and (9),it followsthat(10)In other words,ahigherreduces the primary current over-shoot caused by both the reverse recovery and the interwindingcapacity.However,the lower limitforis .V.E XPERIMENTAL R ESULTSThe outlined concept was implemented and verified on a62.5-kHz converter and the basic circuit diagram is shown in Fig.1.The input-voltagerangenF,TrnF.Whendesigning a transformer for acertaincan be predicted by experimentally (empirically)derivedformulas,and in thiscaseand .The obtained production toleranceof25%.The combined primary series inductance,referred to as primary inductance,is14RHRP8120(Harris),,anddelays the buildup of theprimary current.A common feature of HOV applications is a turn-on spike in the primary current.The origin of that spike is the reverse-recovery characteristic of the high-voltage rectifiers and the interwinding capacitance.A transformer design that minimizes the interwinding capacity either decreases the magnetizing or increases the leakage inductance [8],[9],and the susceptibility of the circuit to ringing is not reduced.An alternative approach to avoid the switching losses associated with the capacitiveGARABANDIC et al.:ZERO-VOLTAGE–ZERO-CURRENT SWITCHING IN PWM CONVERTERS349load is by using the proposed ZVZCS principle.From Fig.11, it is evident that by the time the current turn-on spike builds up,the transistor is already well saturated.In order to correlate the theoretically predicted and experi-mentally obtained performance of the converter,Table I lists several measured and calculated parameters.The efficiency of the62.5-kHz IGBT-based converter versus MOSFET’s(IRFP460)is shown in Fig.12.This plot indicates that the traditional frequency limit for the IGBT’s can be extended to higher frequencies with the use of the proposed ZVZCS.VI.C ONCLUSIONThe interwinding capacitance,which is usually considered undesirable,is in this case constructively used for achiev-ing the ZVZCS.Thus,converters with a high interwinding capacitance(usually HOV converters)can benefit from that parasitic component.The novel ZVZCS method is presented and analyzed,including experimental results obtained from an IGBT-based62.5-kHz dc/dc converter with a rated output voltage of600V and a nominal power of1.2kW.The advantages of the new converter are:•lower switching losses:ZV turn off of the active-to-passive leg transistor,ZVZC turn on of the active-to-passive leg transistor,ZVZC turn off of the passive-to-active leg transistor,and ZC turn on of the passive-to-active leg transistor;•lower noise and reverse-recovery ringing by controlling the charge current of the interwinding capacitance with balancing transformers;•lower conduction losses in the primary since there is no currentflowing during the freewheeling period;•higher power density(IGBT’s can be used at higher switching frequencies).R EFERENCES[1]J.A.Sabate,V.Vlatkovic,R.B.Ridley,F.C.Lee,and B.H.Cho,“Design considerations for high-voltage high-power full-bridge zero-voltage-switching PWM converter,”in IEEE APEC Rec.,1990,pp.275–284.[2]L.H.Mweene,C.A.Wright,and M.F.Schlecht,“A500kHz front-endconverter for a distributed power supply system,”in IEEE APEC Rec., 1989,pp.423–432.[3]J. A.Sabate,V.Vlatkovic,R. B.Ridley,and F. C.Lee,“High-voltage,high-power,ZVS,full-bridge PWM converter employing an active snubber,”in IEEE APEC Rec.,1991,pp.158–163.[4]R.Redl,N.O.Sokal,and L.Balogh,“A novel soft-switching full-bridgedc/dc converter:Analysis,design considerations,and experimental re-sults at1.5kW,100kHz,”in IEEE PESC Rec.,1990,pp.162–172. [5]J.G.Cho,J.A.Sabate,G.Hua,and F.C.Lee,“Zero-voltage andzero-current-switching full bridge PWM converter for high power ap-plications,”in IEEE PESC Rec.,1994,pp.189–194.[6]N.Mohan,T.M.Underland,and W.P.Robbins,Power Electronics:Converters,Applications and Design.New York:Wiley,1989. [7] D.Garabandic,W.G.Dunford,and M.Edmunds,“Snubbers for theinterwinding capacitance in high-output-voltage converters,”in IEEEAPEC Rec.,1998,pp.846–849.[8]S.Ohitsu and T.Ogata,“Considerations on the relationship between dis-tributed capacitance and power loss in a push-pull current-fed converterin buck-mode,”in IEEE PESC Rec.,1988,pp.1007–1012.[9]S.Ohitsu,T.Yamashita,K.Yamamoto,and T.Sugiura,“Stabilityin high-output-voltage push-pull current-fed converters,”IEEE Trans.Power Electron.,vol.8,no.2,pp.135–139,1993.Djordje Garabandic was born in Novi Sad,Yu-goslavia,on December18,1965.He received theB.Sc.degree from the University of Novi Sad,NoviSad,and the Ph.D.degree from the University ofBelgrade,Belgrade,Yugoslavia,in1990and1995,respectively.From1996to1997,he was a Post-DoctoralFellow at the Department of Electrical Engineering,University of British Columbia,Canada.He is cur-rently with the power design team of Xantrex Tech-nology,Inc.,Burnaby,B.C.,Canada.His researchinterests include high-power dc/dc conversion,nondissipative snubbering,and application of the advanced control theory on power conversion systems.He is the author of several technical papers.Dr.Garabandic is the Chair of the IEEE Power Electronics Chapter in Vancouver,Canada.William G.Dunford(SM’92)received the B.Sc.and M.Sc.degrees from Imperial College,London,U.K.,and the Ph.D.degree from the University of Toronto,Toronto,P.Q.,Canada.He spent two years as a Field Engineer with Schlumberger Overseas and has been a Lecturer at Imperial College and the University of Toronto.He is an Associate Professor at the University of British Columbia,Vancouver,Canada. Currently,he is on a leave of absence at Alcatel Espace,Toulouse,France. His engineering interests include efficient and alternative power conversion, particularly at higher powers.He also has projects in battery management and modeling.Dr.Dunford currently chairs the Chapter Development for the IEEE Power Electronics Society and was the General Chairman of PESC in1986(and will be in2001).Mark Edmunds,photograph and biography not available at the time of publication.。

加钳位二极管的零电压开关PWM三电平直流变换器Z ero 2Voltage 2Switching PWM Three 2Level Converter with Clamping Diodes阮新波　许大宇　严仰光(南京航空航天大学　210016)R u an X inbo X u Dayu Y an Y anggu ang (Nanjing University of Aeronautics &Astronautics 210016　China ) 摘要　零电压开关PWM 三电平直流变换器(ZVS PWM TL 变换器)利用变压器的漏感和开关管的结电容可以实现开关管的零电压开关,但是输出整流管仍然存在反向恢复带来的尖峰电压。

为了解决这个问题,提出一种新的ZVS PWM TL 变换器,它在基本的ZVS PWM TL 变换器中增加两个二极管,消除了输出整流管的电压振荡,同时保留基本的ZVS PWM TL 变换器的所有优点。

分析了这种新的变换器的工作原理,并在一个600W 的原理样机上进行了验证,最后给出了实验结果。

关键词:三电平直流变换器　零电压开关　脉宽调制中图分类号:TM463Abstract Zero 2Voltage 2Switching PWM Three 2Level Converter (ZVS PWM TL Converter )can realize ZVS for the power switches with the use of the leakage inductance of the transformer and the output capacitors of the power switches.However ,the output rectifier diodes still suffer the volt 2age spike and voltage oscillation resulted by the reverse recovery of the rectifier diodes.In this paper a novel ZVS PWM TL converter is proposed which introduces two clamping diodes to the basic ZVS PWM TL converter ,thus the voltage oscillation and voltage spike are eliminated.The operation prin 2ciple of the novel converter is analyzed and verified by a 600W prototype converter.The experimental results are also included in this paper.K eyw ords :Three 2level converter Zero 2voltage 2switching Pulse 2width 2modulation中国航空基础科学基金和台达电力电子科教发展基金资助项目。

国 家 自 然 科 学 基 金 （ 51077112 ） 和 河 北 省 自 然 科 学 基 金 （E2015203407，13211907D-2）资助项目。 收稿日期 2016-07-06 改稿日期 2016-08-17
（超级电容或蓄电池）需要双向 DC-DC 变换器实 现功率双向传输[3,4]。在双向 DC-DC 变换器中，非 隔离型 Buck-Boost 双向变换器因为具有结构简单、 可靠性高、成本低等优点而备受青睐[5-8]。然而，在 电动汽车这种高电压、大功率应用场合中需要选用 耐压高的开关管。开关管的导通阻抗和寄生电容随 耐压值的升高而增大，影响变换器效率。文献[9-12] 提出一种非隔离型单向三电平 Buck-Boost 变换器， 可降低开关管的电压应力，且为输入电压的一半。
孙孝峰 袁 野 王宝诚 李 昕 潘 尧
（电力电子节能与传动控制河北省重点实验室（燕山大学） 秦皇岛 066004）
摘要 针对非隔离型三电平 Buck-Boost 双向变换器，提出一种零电压开通（ZVS）实现方案。 该方案在不添加任何辅助元件的情况下，可使非隔离型三电平 Buck-Boost 变换器的所有开关管在 全负载范围内实现 ZVS，提高变换器的效率。此外，利用异相控制、电感电流倍频降低电感的体 积，提高功率密度。首先对实现 ZVS 的工作过程进行分析，并且分析反向电流 IR 对软开关的影 响；然后推导出死区时间和开关频率表达式；最后搭建实验样机，通过 Buck 模式和 Boost 模式的 实验来验证该方案的正确性和有效性。
Keywords：Bidirectional converter, zero-voltage switching, inductor current frequency doubliHale Waihona Puke g, reverse current

一种新颖的同步BUCK芯片过零检测电路的设计黄苏平;冯全源【摘要】同步BUCK变换器在轻载模式下会出现电感电流倒灌现象,这会使得芯片的效率大大降低。

针对这一问题,设计了一款过零检测电路,该电路利用MOS管工作在线性区时的沟道电阻来产生过零检测比较器的负阈值电压,从而有效地限制了电流的倒灌；还设计了边沿隐匿电路,避免电路切换时引起的误触发。

该过零检测电路基于0.5μm BCD工艺,在HSPICE软件上仿真验证,当系统温度在0~85℃变化时,负阈值容差为10 mV,过零检测电路性能良好。

%The phenomenon that inductor current flow backward appears when synchronous BUCK converter works in light load mode,and the efficiency of the system will be reduced. In response to this problem,a novel zero-cross detection circuit is designed. By using the channel resistance when the MOS transistor is working in linear region to realize negative threshold voltage for zero-cross detection comparator,the happening of current backward flowing is avoided. Edge occult circuit is also designed to avoid false triggering at switching moment. The zero-cross detection circuit is based on 0. 5 μm BCD technology,and has been carried on HSPICE. The simulation result shows that the negative threshold tolerance is only 9 mV when system ranges in temperature from -25 ℃ to 50 ℃,and the zero-cross detection circuit works well.【期刊名称】《电子器件》【年(卷),期】2014(000)003【总页数】4页(P408-411)【关键词】BUCK变换器;轻载模式;过零检测;同步【作者】黄苏平;冯全源【作者单位】西南交通大学微电子研究所，成都610031;西南交通大学微电子研究所，成都610031【正文语种】中文【中图分类】TN433近年来，随着低电压、大电流、高效率开关电源的越来越广泛应用，采用低功耗的同步整流器替代传统的整流二极管已成为提高效率的主要技术［1-3］。