南京航空航天大学金城学院
毕业设计（论文）外文文献翻译
系部自动化
专业电气工程与自动化
学生姓名周治世学号2012031117
指导教师曹鑫职称副教授
2016年2月
Elimination of Mutual Flux Effect on Rotor Position
Estimation of Switched Reluctance Motor Drives
Jin Ye, Student Member, IEEE, Berker Bilgin, Member, IEEE, and Ali Emadi, Fellow, IEEE
Abstract—An approach to eliminate mutual flux effect on rotor position estimation of switched reluctance motor drives at rotating shaft conditions without a prior knowledge of mutual flux is proposed in this paper. Neglecting the magnetic saturation, the operation of conventional self-inductance estimation using phase current slope difference method can be classified into three modes: Mode I, II, and III. At positive-current-slope and negative-current-slope sampling point of one phase, the sign of current slope of the other phase changes in Mode I and II, but does not change in Mode III. Theoretically, based on characteristics of a 2.3 kW, 6000 rpm, three-phase 12/8 SRM, mutual flux introduces a maximum ±7% self-inductance estimation error in Mode I and II, while, in Mode III, mutual flux effect does not exist. Therefore, in order to ensure that self-inductance estimation is working in Mode III exclusively, two methods are proposed: variable-hysteresis-band current control for the incoming phase and variable-sampling self-inductance estimation for the outgoing phase .Compared with the conventional method which neglects mutual flux effect, the proposed position estimation method demonstrates an improvement in position estimation accuracy by 2?. The simulations and experiments with the studied motor validate the effectiveness of the proposed method.
Manuscript received February 6, 2014; accepted April 14, 2014. Date of publication April 24, 2014; date of current version October 15, 2014. This work was supported in part by the Canada Excellence Research Chairs Program. Recommended for publication by Associate Editor R. Kennel.
The authors are with the McMaster Institute for Automotive Research and Technology, McMaster University, Hamilton, ON L8S 4K1 Canada (e-mail: yej25@mcmaster.ca; bilgin@mcmaster.ca; emadi@mcmaster.ca).
Color versions of one or more of the figures in this paper are available onlineat https://www.doczj.com/doc/973464003.html,.
Digital Object Identifier 10.1109/TPEL.2014.2319238
I. INTRODUCTION
S WITCHED reluctance motor (SRM) is a cost effective solution for the automotive industry due to absence of rotor windings and permanent magnet on the rotor [1]–[9]. In general, the encoder or resolver is installed to obtain the rotor position and speed for the torque or speed control of SRM. This increases the cost and volume of the motor drive, and reduces the reliability. Position sensorle- ss control of SRM is widely studied in the literature [10]–[29]. Magnetic characteristics of the SRM
including the flux, self-inductance, and back electromagnetic force (EMF) are rotor position depen- dent, and therefore, these parameters can be estimated to obtain the rotor position. Pulse injection method [16]–[20] is one of the rotor positions sensorless methods for low speed operation. High-frequency signal is injected to the inactive phase to obtain inductance, which is later converted to rotor position. In [16], the working sector of the SRM is selected by comparing amplitude of current response of inactive phases with two predefined thresholds. This method is simple but it is not promising for instantaneous torque control, where real-time rotor position is required.
In [17], voltage pulse is injected to determine the speed range of SRM and then the corresponding dynamic model is defined for rotor position estimation. In [18], a rotor position estimation scheme based on phase inductance vector is proposed. The pulse is injected to get the inductance of inactive phases and full cycle inductance is obtained. This method has advantages of no need of a priori knowledge on magnetic characteristics of the machine. Some pulse injection methods with an additional circuit [20] are also reported and this may increase the cost and implementation complexity. In summary, voltage injection methods often suffer from either additional power losses or low speed constraint. Computation intensive methods such as observer based estimation, and neural network are also presented in [21]–[26]. Although they show robustness or model in dependence, they are complicated and not promising solutions for practical applications.
Passive rotor position estimation based on measurement of terminal voltage and phase current of active phases are gaining interest due to absence of additional hardware or power losses .One of these methods is to estimate the flux linkage of SRM. A simple approach to
obtain the flux linkage is to use the integration of the terminal voltage subtracted by the voltage across the ohmic resistance. This method shows poor accuracy at low speed when back EMF is small. Also, the accuracy is deteriorated by variation of the ohmic resistance and accumulation error due to integration. Several methods are proposed to improve the accuracy of the estimation. In [27], the flux linkage variation instead of flux linkage is used to estimate the rotor position and error analysis of the rotor position estimation is provided. In [28], flux linkage is estimated based on the amplitude of the first-order
switching harmonics of the phase voltage and current. The influence of resistance variation and low back EMF for rotor position estimation is suppressed, and therefore, both accuracy and applicable speed ranges are improved. Self-inductance-based rotor position estimation [18], [29], [30] is an alternative approach of passive position sensorless techniques. By neglecting variation of the speed, back EMF and ohmic resistance in a switching period, the self-inductance is estimated by measuring the phase current slope difference [18]. In [30], incremental inductance is estimated by using phase current slope difference and therefore the application of this method is extended to magnetic saturated region. Compared to flux linkage methods, the influence of the variation of resistance is eliminated and it is capable of operating at low speeds. However, as the speed increases, the overlapping region of the active phases becomes significant and mutual flux cannot be neglected anymore. The accuracy of both inductances based and flux linkage-based rotor position estimation methods are decreased at higher speed due to mutual flux between active phases. In addition, torque sharing function (TSF) [31], [32] is widely used in instantaneous torque control to reduce commutation torque ripples.When TSF is applied, overlapping areas of incoming and outgoing phases are significant even at low speed. Therefore, mutual flux has to be considered to achieve accurate estimation of rotor position over a wide speed range. Although the influence of the mutual flux on modeling of the SRM is investigated in [33]–[35], publications about the effect of the mutual flux on rotor position estimation are still limited. In [36], mutual flux of SRM with even and odd number of phases is studied and the effect of the mutual flux on position estimation is verified by both simulation and experiments. By calibrating the flux linkage with the measured mutual flux by experiments, the accuracy of position estimation is increased by 3o. In [37], mutual flux linkage obtained
by finite element analysis (FEA) is applied to compensate the observed flux linkage. Accuracy of the rotor position estimation is proved to be increased. The mutual flux needs to be either measured or simulated in advance in the two methods proposed in [36] and [37]. However, this process is time consuming. Due to the measurement noise or manufacturing imperfections, mutual flux may not be precise. Meanwhile, flux-linkage based rotor position estimation method is still not desirable at low speed.
In this paper, two methods to eliminate mutual flux effect on rotor position estimation are presented, which do not require a priori knowledge of mutual flux linkage profiles of SRM. In order to investigate the mutual flux effect on rotor position estimation, a dynamic model of SRM incorporating mutual flux is obtained. Then, the self-inductance estimation error due to mutual flux is theoretically derived by using phase current slope difference method. Three operational modes are defined during self-inductance estimation. In Modes I and II, at the positive-current-slope and negative-current-slope sampling point of a phase, the sign of the current slope of the other phase changes. In Mode III, the sign of the current slope of the other phase does not change. Based on magnetic characteristics of the studied SRM, in Modes I and II, the mutual flux introduces a maximum ±7% self-inductance estimation error. However, in Mode III, the impact of mutual flux on estimation of self-inductances does not exist. Therefore, variable-hysteresis band current controller and variable-sampling
self-inductance estimation methods are proposed, so that the self-inductance estimation operates in Mode III exclusively and, hence, a priori knowledge of mutual flux becomes unnecessary. By applying variable-hysteresis band current controller method in the incoming-phase self-inductance estimation, the switching state of the outgoing phase is kept unchanged and incoming-phase self-inductance estimation is forced to work in Mode III. The hysteresis-band of the outgoing phase is adjusted accordingly. When applied in the outgoing phase self-inductance estimation, variable-hysteresis band current controller introduces undesirable higher ripples. For this reason, variable-sampling self-inductance estimation is proposed for the outgoing-phase self-inductance estimation During the outgoing-phase self-inductance estimation; the negative-current-slope sampling point is adjusted to ensure that the current slope of incoming-phase has the same sign at the positive and negative current-slope sampling point of the outgoing phase. Finally, phase
self-inductance estimation in the active region for each phase is introduced. Simulation and experimental results are provided to verify the performance of the proposed rotor position estimation scheme at rotating shaft conditions.
II. DYNAMIC MODEL OF SRM CONSIDERING MUTUAL FLUX
A. Circuit Modeling of SRM With Mutual Flux
In a three-phase SRM, no more than two phases are conducted simultaneously. During commutation, incoming and outgoing phases are denoted as k th and (k–1) th phases, respectively. Phase voltage equations are derived as (1)
v k=Ri k+eλk et
V K?1=Ri k?1+eλk?1
et
1
where v k, i k, and λk are the phase voltage, current, and flux linkage of k th phase, respectively; v k?1, i k ?1, and λk?1 are the phase voltage, current, and flux linkage of (k–1)th phase, respectively.
As the speed of SRM increases, overlapping areas of the two phases are increased significantly due to higher back EMF. This causes the effect of mutual flux to increase and it cannot be neglected anymore. When mutual flux is considered, flux linkage for incoming and outgoing phases can be expressed as (2)
λk=λk,k+λk,k?1
λk?1=λk?1,k?1+λk?1,k (2)
where λk,k and λk?1,k?1 are the self-flux linkages of k th and (k–1)th phase; λk,k?1 and λk?1,k are mutual flux linkages.
Neglecting the magnetic saturation, the flux linkage is a linear function of the inductance and, hence, (2) can be reorganized as (3)
λκλκ?1=
L k,k M k,k?1
M k?1,k L K?1,K?1
i k
i k?1 (3)
where L k,k and L k?1,k?1 are the self-inductances of the k th and (k–1)th phase; M k,k ?1 and M k?1,k
are the mutual inductances.
The mutual inductance between two conducted phases meets (4)
M k,k?1=M k?1,k (4)
Fig.1. FEA inductance and torque profiles of 12/8 SRM. (a) Inductance profiles. (b) Torque profiles. Substituting (3) for (1) and (2), the phase voltage equations are derived as (5)
v k
v k?1=R
i k
i k?1+
L k,k M k,k?1
M K?1,K L k?1,k?1
di k
dt
di k?1
dt
+ωm
?L k,k
?θ
?M k,k?1
?θ
?M k?1,k
?θ
?L k?1,k?1
?θ
i k
i k?1 (5)
where θ and ωm are rotor position and angular speed of SRM.
Electromagnetic torque of k th phase is represented as (6) neglecting magnetic saturation.
T kθ,i=1
2
?Lθ,i k
?θ
i k26
Where T k is the torque produced by k th phase, and i k is the k th phase current.
For an n-phase SRM, total electromagnetic torque T is rep- resented as (7)
T=T k
n
k=1
(7)
B. Analysis of Mutual Flux of SRM
The FEA of the studied SRM is conducted in JMAG software [38] and the nonlinear inductance profile and torque profile of studied SRM is shown in Fig. 1(a) and (b), respectively. In three-phase SRM, two phases are excited during commutation The magnetic flux density distribution of 12/8 SRM during one phase and two-phase excitation are shown in Fig. 2(a) and (b), respectively. Compared with one-phase excitation mode, the two-phase excitation works at short-flux path [33] and the flux linkage of an individual phase includes both self and mutual flux linkage.
Fig.2. Magnetic flux density of a 12/8 SRM. (a). Single phase excitation. (b) Two-phase excitation during phase commutation.
Due to alternate polarities of windings of a three-phase motor, mutual flux is always additive and symmetric among individual phases. For the same current on adjacent phases, the mutual inductance profiles obtained from FEA are shown in Fig. 3(a). MA, B is the mutual inductance between phases A and B. The maximum value of mutual inductance is around 2% of the self- inductance at the same current level. The spatial relationship between self-inductance and mutual inductance of 12/8 SRM is also shown in Fig. 3. The mutual inductance profile MA, B is shifted by around 7.5?compared with the self-inductance of
phase A, LA.
Self-inductance.
Fig.4. Hysteresis control of phase current of SRM.
III. ERROR ANALYSIS OF SELF-INDUCTANCE ESTIMATION DUE TO MUTUAL
FLUX
A. Self-Inductance Estimation Without Considering the Mutual Flux
Hysteresis controller is applied for phase current control, as shown in Fig. 4. Upper and lower current references of the k th phase are denoted as i k up and i k_low, respectively. So, the hysteresis band is represented as (8)
Δi k=i k?up?i k?low (8) When the switches T1 and T2 are turned ON as shown as Fig. 5(a), dc-link voltage is applied and the phase current slope is positive. When the switches T1 and T2 are turned OFF, shown as Fig. 5(b), the phase current slope is negative.
The voltage equations neglecting magnetic saturation are then derived as (9) and (10) when switches are ON and OFF, respectively.
Fig.5. Two switching states of SRM driver. (a) ON state. (b) OFF state.
U dc
=Ri k+L k,k di k(t k?on)
dt
+
?L k,k
eθ
i kωm (9)
?U dc=Ri k+L k,k di k(t k?off)
dt +?L k,k
eθ
i kωm
(10)
where t k?on and t k?off are time instant when the k th phase switching states are ON and OFF
during a switching period, respectively; di k(t k?on)
dt and di k(t k?off)
dt
are the slope of k th phase
current at t k?on and t k?off, respectively; and U dc is the dc-link voltage.
The switching period is short enough and, therefore, variation of the mechanical speed, inductance, back EMF and resistance are neglected. The self-inductance can be derived as (11) by combining (9) and (10). For a given dc-link voltage, unsaturated self-inductance can be estimated by using the phase current slope difference between ON and OFF states.
L k,k =2U dc
k k ?on dt ?k k ?off
dt (11)
where L k,k is estimated k th phase self-inductance without considering the mutual flux.
B. Analysis of Self-Inductance Estimation Error Due to Mutual Flux
In the previous section, self-inductance estimation is derived by neglecting the mutual flux between two adjacent phases. When overlapping areas of two phases can be neglected, self- inductance estimation without considering the mutual flux is ac- curate. As the speed increases, the overlapping region becomes significant and the mutual inductance cannot be neglected. Con- sidering the mutual inductance, the k th phase voltage equation is derived as
(12) and (13) when k th phase switches are ON state and OFF state, respectively
U dc =Ri k +L k,k
di k t k ?on dt +?L k,k eθi k ωm
+ M k,k ?1di k ?1(t k ?on )dt
+?M k,k ?1eθi k ?1ωm （12）
Fig.6. Illustration of three modes during self-inductance estimation of k th phase.
?U dc =Ri k +L k,k di k t k ?off dt +?L k,k eθi k ωm
+ M k,k ?1di k ?1(t k ?off )dt
+?M k,k ?1eθ
i k ?1ωm (13) where di k t k?on dt and di k t k?off dt
are the slopes of k th phase current at t k?on and t k?off , respectively.
Similarly, the variation of the mechanical speed, inductance, Similarly, the variation of the
mechanical speed, inductance, L k,k?m
=2U dc?M k,k?1
di k?1t k?on
dt?
di k?1t k?off
dt
di k t k?on
dt?
di k t k?off
dt
14
where L k,k-m is the estimated k th phase self-inductance considering mutual flux. The error of self-inductance estimation due to mutual flux is derived as (15)
err k=L k,k?m?L k,k
L k,k?m
=
?M k,k?1di k?1t k?on
dt
?di k?1t k?off
dt
2U dc?M k,k?1di k?1t k?on
dt
?di k?1t k?off
dt
(15)
Where errk is k th phase self-inductance estimation error due to the mutual flux from (k–1)th phase.
In order to analyze the self-inductance estimation error due to mutual flux, three modes are defined during k th phase self-inductance estimation as shown in Fig. 6: Mode I, II and III. Since the self-inductance of the incoming phase (k th phase) is much lower, k th phase current slope is much higher than (k–1) th phase. Upper and lower current references of k th phase are denoted as i k?up and i k?low, and upper and lower current references of (k–1)th phase are denoted as i k?up and i k?low. The positive-current-slope and negative-current-slope of k th phase is sampled at t k?on(III)and t k?off(III)in Mode III, t k?on(II)and t k?off(II)in Mode II, and t k?on(I)and t k?off(I)in Mode I, respectively. The self-inductance estimation error due to mutual flux in Mode I, II and III is derived below. In Mode I and Mode II, the mutual flux leads to self-inductance estimation error, while in Mode III, the mutual flux effect is negligible.
1) Self-Inductance Estimation Error in Mode I: In Mode I, at positive-current-slope sampling point t k?on(I)and negative current-slope sampling point t k?off(I)of k th phase, (k–1)th phase current slope is positive and negative, respectively. Considering the mutual flux from k th phase, the (k–1) th phase voltage equation can be derived as (16) and (17) at t k?off(I) and t k?off(I)
U dc=Ri k?1+L k,k?1di k?1t k?on I
dt
+?L k?1,k?1
eθ
i k?1ωm
+M k,k?1di k(t
k?on（I）
)
dt
+?M k,k?1
eθ
i kωm（16）
?U dc=Ri k?1+L k,k?1di k?1t k?off I
dt
+?L k?1,k?1
eθ
i k?1ωm
+M k,k?1di k(t
k?off（I）
)
dt
+?M k,k?1
eθ
i kωm（17）
Subtracting (17) from (16), (18) can be derived.
L k?1,k?1di k?1t
k?on（I）
dt
?
di k?1t k?off（I）
dt
+M k,k?1di k t k?on I
dt
?
di k t k?off I
dt
=2U dc18 Similarly, subtracting (12) by (13), (19) can be derived for k th phase
L k,k di k t
k?on（I）
dt
?
di k t k?off（I）
dt
+M k,k?1di k?1t
k?on（I）
dt
?
di k?1t
k?off（I）
dt
=2U dc (19)
Subtracting (18) from (19), (20) can be derived
di k t k?on I
dt ?di k t k?off I
dt
=L k?1,k?1?M k,k?1
L k,k?M k,k?1di k t k?on I
dt
?
di k t k?off I
dt
（20）
Substituting (20) for (18), the (k–1) th phase current slope difference considering mutual flux from k th phase in Mode I is derived as (21)
di k t k?on I
dt ?
di k t k?off I
dt
=
2U dc
L k ?1,k ?1+M k,k ?1L k ?1,k ?1?M k,k ?1
L k,k ?M k,k ?1 （21）
Substituting (21) for (15), the error of k th phase self- inductance estimation due to mutual flux from (k –1)th phase in Mode I is calculated as (22)
err k(I)=?M k,k ?1L k ?1,k ?1+M k,k ?1L k ?1,k ?1?M k,k ?1L k,k ?M k,k ?1
?M k,k ?1 (22) Where errk (I) is k th phase self-inductance estimation error due to mutual flux from (k –1)th phase in Mode I.
2) Self-Inductance Estimation Error in Mode II: In Mode II, at positive-current-slope sampling point t k on (II) and negative-current-slope sampling point t k o ? (II) of k th phase, (k –1) th phase current slope is negative and positive, respec- tively. Similarly, in Mode II,
(23) is derived for the (k –1) th phase
L k ?1,k ?1 di k ?1 t k ?off （II ） dt ?di k ?1 t k ?on （II ） dt
+M k,k ?1 di k t k ?on II dt ?di k t k ?off II dt
=2U dc 23
By using the same approach as Mode I, the error of k th phase self-inductance estimation due to mutual flux from (k –1) th phase in Mode II is calculated as (24)
err k(II)
=M k,k ?1L k ?1,k ?1+M k,k ?1L k ?1,k ?1?M k,k ?1L k,k ?M k,k ?1
+M k,k ?1 (24) where errk (II) is k th phase self-inductance estimation error due to mutual flux from (k –1) th phase in Mode II.
3) Self-Inductance Estimation Error in Mode III: In Mode III, at positive-current-slope sampling point t k?on III and negative-current-slope sampling point t k?off III of k th phase, (k –1) th phase current slope has the same sign (either both neg- ative or positive). By neglecting the inductance and back EMF variation, (25) is derived. Substituting (25) for (15), error of k th phase self-inductance estimation due to mutual flux from (k –1) th phase in Mode
III can be calculated as (26). The self- inductance estimation error is zero, and therefore, the mutual flux coupling effect on self-inductance estimation is eliminated
di k?1t k?on III
dt ?
di k?1t k?off III
dt
=0 （25）
err k(III)
=0 （26）
where errm (III) is k th phase self-inductance estimation error due to mutual flux from (k–1) th phase in Mode III.
The error analysis of the outgoing phase (k–1) th phase self- inductance estimation in Mode I, II, and III can also be derived following the same procedure above. Based on the magnetic characteristics of the studied SRM shown in Fig. 3, the absolute values of self-inductance estimation error of phase A due to mutual flux from phase B and phase C in Mode I and II are shown in Fig. 7. Phase A self-inductance estimation error due to mutual flux from phase B and phase C is rotor position dependent. The mutual flux introduces maximum around 7% and minimum around 1% error to the phase self- inductance estimation in Mode I and II. In Mode III, the mutual flux introduces 0% self- inductance estimation error. Therefore, the methods to ensure the self-inductance estimation working in Mode III exclusively are proposed and explained in the next section.
Fig.7. Self-inductance estimation error of phase A due to mutual flux from phase B and C.
IV. PROPOSED SELF-INDUCTANCE ESTIMATION TO ELIMINATE
MUTUAL FLUX EFFECT
Based on the magnetic characteristics of the studied motor, the mutual flux introduces a maximum ±7% self-inductance estimation error in Mode I and II, while the mutual flux effect does not exist in Mode III. The proposed technique here is based on excluding Modes I and II and, hence, eliminating the error in self-inductance estimation due to the mutual flux. Two methods are proposed which utilizes the operation of Mode III exclusively for self-inductance estimation: variable-hysteresis-band current control for the incoming phase and variable-sampling method for the outgoing phase. In the variable-hysteresis-band current control method, when estimating the phase self-inductance of the incoming phase, the variation in the switching states of the other phase is avoided and the hysteresis band of the outgoing phase is adjusted. However, when the variable-hysteresis-band current control is applied to the outgoing-phase self-inductance estimation, undesirable higher current ripples might be observed in the incoming phase. For this reason, variable-sampling method for the outgoing-phase self-inductance estimation is proposed to overcome the drawback of variable-hysteresis-band current controller when applied to the outgoing-phase.
A. Variable-Hysteresis-Band Current Control for Incoming Phase
1) Principle of the Proposed Method for Self-Inductance Estimation of kth Phase (Incoming Phase): Fig. 8 illustrates the principle of the proposed variable-hysteresis-band current control during the k th phase self-inductance estimation.
During commutation, the k th phase current slope is much higher than that of (k–1)th because of lower self-inductance of k th phase. The (k–1) th phase current profiles with constant- hysteresis-band control and proposed variable-hysteresis-band control are denoted as solid and dotted line, respectively. The basic concept of the variable-hysteresis-band current control method is to make sure that the switching state of (k–1) th phase is unchanged during the time intervals t k?on(II)?t k?off(II)and t k?on(I)?t k?off(I). Therefore, the sign of (k–1) th phase current slope is unchanged in these intervals. When the
self-inductance estimation of k th phase is completed at t k ?off (II)and t k ?off (I), switches of (k –1) th phase are turned OFF or ON according to the error between (k –1) th phases current and its reference. When (k –1) th phase current at t k ?off (I)or t k ?off (II)is lower than its lower reference i k ?1_low , switches are turned ON. When (k – 1) th phase current t k ?off (I) or t k ?on (II) is higher than it’s upper .
Fig.8. Illustration of the proposed variable-hysteresis-band current controller.
Hysteresis band for k th phase current control (Δik) stays con- stant. Its upper and lower reference is denoted as i k-up and i k-low , respectively. However, in order to keep the switching state of (k –1) th phase unchanged during the k th phase self-inductance estimation, the hysteresis-band of (k –1) th phase (Δi k?1) varies with time. As shown in Fig. 8, the hysteresis band of (k –1) th phase is represented as (27). Δi k ?1= i k ?1_up +Δi k ?1 I
? i k ?1low ??i k ?1 II (27)
where Δi k?1(I) and Δi k?1(II) are the adjusted hysteresis band of (k –1)th phase current in Mode I and II during k th phase self-inductance estimation, respectively.
2)Analysis of Adjusted Hysteresis Band: The proposed method increases the hysteresis band and it introduces more current ripples. Therefore, it is necessary to analyze the adjusted hysteresis band for (k –1)th phase. The adjusted hysteresis band of Δi k?1(I) and Δi k?1(II) varies with time. At time instants t I and t II , the (k –1) th phase current reaches its upper and lower reference. Neglecting the mutual flux from k th phase, the1) th phase voltages during
the
intervals t I?t k_off(I)are derived as (28)
U dc=Ri k?1+L k?1,k?1
?i k?1(I)
t k?off I?t I
+
?L k?1,k?1
?θ
i k?1ωm (28)
The adjusted hysteresis band for Mode I are derived as (29) according to (28) ?i k?1(I)
= U dc?Ri k?1?
?L k?1,k?1
?θi k?1ωm t k?off I?t I
L k?1,k?1
（29）
In order to obtain the range of the adjusted hysteresis band, the range of back EMF should be obtained firstly. In current control mode, the back EMF of SRM cannot exceed the dc-link voltage, (30) has to be satisfied
?U dc≤?L k?1,k?1
?θ
i k?1ωm
≤U dc（30）
Considering the maximum possible back-EMF in (30) and the resistive voltage drop, (31) has to be satisfied
U dc?Ri k?1??L k?1,k?1
?θ
i k?1ωm≤U dc?
?L k?1,k?1
?θ
i k?1ωm
≤U dc+U dc31
In addition, as shown in Fig. 8, time intervals have to meet the constraints (32) t k?off I?t I
≤t k?off I?t k?on I（32）Based on (29), (31) and (32), the adjusted hysteresis band for Mode I must satisfy (33
?i k?1(I)=（U dc?Ri k?1?
?L k?1,k?1
?θi k?1ωm）（t k?off I?t I）
L k?1,k?1
≤2U dc t k?off I?t k?on I
L k?1,k?1
33
?i k?1(I)
max
=
2U dc t sample
L k?1,k?1
(34)
where the required sample time is represented as
t sample=t k?off I?t k?on I
Similarly, the maximum adjusted hysteresis band in Mode II ?i k?1(I)
max can be derived, and it has the same expression with (34). With the same sampling time t sample in Mode I and II, the maximum adjusted hysteresis band in Mode I and Mode II is the same. The maximum adjusted hysteresis band is a function of self-inductance. During commutation, the (k–1)th phase (outgoing phase) L k?1,k?1 is close to aligned inductance La. Therefore, the maximum adjusted hysteresis band is approximated
as (35). The aligned inductance is relatively large in SRM, and, therefore, only a slight increase in current hysteresis band and current ripple can be expected.
?i k?1(I)
max
=2U dc t sample
La
（35）
3) Drawback of The Proposed Method For Self-Inductance Estimation of (k–1) th Phase (Outgoing Phase): Fig. 9 illustrates the principle of k th phase variable-hysteresis-band current control during (k–1) th phase self-inductance estimation. During commutation, the k th phase self-inductance L k,k is close to unaligned inductance Lu and the maximum adjusted hysteresis band is represented as (36)
?i k max
=2U dc t sample
L u
（36）
开关磁阻电机工作原理及其驱动系统 开关磁阻电机 Switched Reluctance Drivesystem, SRD 开关磁阻电机驱动系统(Switched Reluctance Drive system, SRD)具有一些很有特色的优点:电机结构简单、坚固、维护方便甚至免维护，起动及低速时转矩大、电流小;高速恒功率区范围宽、性能好，在宽广转速和功率范围内都具有高输出和高效率而且有很好的容错能力。这使得SR电机驱动系统在家用电器、通用工业、伺服与调速系统、牵引电机、高转速电机、航空航天等领域得到广泛应用。 SR电机是一种机电能量转换装置。根据可逆原理，SR电机和传统电机一样，它既可将电能转换为机械能——电动运行，在这方面的理论趋于成熟;也可将机械能转换为电能——发电运行，其内部的能量转换关系不能简单看成是SR电动机的逆过程。 开关磁阻电机的发展概况和发展趋势 “开关磁阻电机(Switched reluctance motor)”一词源见于美国学者 S.A.Nasarl969年所撰论文，它描述了这种电机的两个基本特征：①开关性——电机必须工作在一种连续的开关模式，这是为什么在各种新型功率半导体器件可以获得后这种电机才得以发展的主要原因；②磁阻性——它是真正的磁阻电机，定、转子具有可变磁阻磁路，更确切地说，是一种双凸极电机。开关磁阻电机的概念实际非常久远，可以追溯到19世纪称为“电磁发动机”的发明，这也是现代步进电机的先驱。在美国，这种电机常常被称为“可变磁阻电机(variable reluctance motor, VR电机)”一词, 但是VR电机也是步进电机的一种形式，容易引起混淆。有时人们也用“无刷磁阻电机(Brushless reluctance motor)”一词，以强调这种电机的无刷性。“电子换向磁阻电机(Electronically commutated reluctance motor)”一词也曾采用，从工作原理来看，甚至比“开关磁阻”的说法更准确—些，但也容易与电子换向的水磁直流电机相混淆。毫无疑问，正是由于英国 P．J．Lawrenson教授及其同事们的杰出贡献，赋予了现代SR电机新的意义，开关磁阻电机一词也因此逐渐为人们所接受和采用。 从电机结构和运行原理上看，SR电机与大步距角的反应式步进电机十分相似，因此有人将SR电机看成是一种高速大步距角的步进电机。但事实上，两者是有本质差别的，这种差别体现在电机设计、控制方法、性能特性和应用场合等方面，见表11-1。
Journal of Electrical Engineering 电气工程, 2016, 4(1), 55-62 Published Online March 2016 in Hans. https://www.doczj.com/doc/973464003.html,/journal/jee https://www.doczj.com/doc/973464003.html,/10.12677/jee.2016.41008 Speed Control Strategy of Switched Reluctance Motor Zhou Du1,2, Dingxiang Wu2,3, Lijun Tang1,2 1School of Physics and Electronic Sciences, Changsha University of Science & Technology, Changsha Hunan 2Hunan Province Higher Education Key Laboratory of Modeling and Monitoring on the Near-Earth Eletromagnetic Environments, Changsha Hunan 3Billion Set Electronic Technology Co, Ltd., Changsha Hunan Received: Mar. 1st, 2016; accepted: Mar. 19th, 2016; published: Mar. 24th, 2016 Copyright ? 2016 by authors and Hans Publishers Inc. This work is licensed under the Creative Commons Attribution International License (CC BY). https://www.doczj.com/doc/973464003.html,/licenses/by/4.0/ Abstract Aimed at research on starting mode and speed control of switched reluctance motor speed control system, a two-phase starting is adopted to start the electric, in order to increase the torque and reduce the torque ripple. A fuzzy adaptive PID control algorithm is proposed, and a switched re-luctance motor speed control system with STM32 + FPGA as the main controller is designed, ap-plying current chopping in low speed and angle position control mode in high speed, which has a certain effect on solving the problems of high overshoot, slow dynamic response and low accuracy. The experimental results show that the precision of the system speed is within 10 r/min, and the maximum overshoot is 15 r/min. Keywords Switched Reluctance Motor, Torque Ripple, Fuzzy Adaptive Tuning PID 开关磁阻电机速度控制 杜舟1,2，吴定祥2,3，唐立军1,2 1长沙理工大学物理与电子科学学院，湖南长沙 2近地空间电磁环境监测与建模湖南省普通高校重点实验室，湖南长沙 3长沙亿旭机电科技有限公司，湖南长沙
题目：开关磁阻电机
开关磁阻电机 学习《特种电机及其控制》这门课程，这要介绍了无刷直流电机及其控制、开关磁阻电机及其控制系统、步进电机及其控制，其中我最感兴趣的开关磁阻电机。下面我将对我所了解的开关磁阻电机做一总结。 一、发展背景 开关磁阻电机是80年代初随着电力电子、微电脑和控制技术的猛烈发展而发展起来的一种新型调速驱动系统，具有结构简单、运行可靠及效率高等突出优点，成为直流电机调速系统、交流电机调速系统和无刷直流电机调速系统强有力的竞争者，引起各国学者和企业界的广泛关注，目前开关磁阻电机已开始应用于工业、航空业和家用电器等各个领域。 开关磁阻电机的基本概念可追溯到19世纪40年代，1842年，英国的Aberdeen和Dafidson用两个U型电磁铁制造了由蓄电池供电的机车电动机。20世纪60年代，大功率晶闸管的出现为SR电机的研究发展提供了重要的物质条件。1967年，英国的Leeds大学开始对SR电机进行深入研究；直到1970年左右，研究结果表明：SR电机可以在单相电流下四象限运行，功率变换器无论是用晶体管还是用普通晶闸管，所需开关数都是最少的；电动机成本也明显低于同容量的感应电动机。20年代70年代初，美国福特公司研制出最早开关磁阻电机的调速系统，其结构为轴向气隙电动机，具有电动机和发电机运行状态和较宽范围调速的能力，适合于蓄电池供电的电动车辆的转动。1980年Leeds大学的Lawrenson教授及其同事总结出了自己的研究成果，发表了题为“Variable--Speed Switched Reluctance Motors”的论文，系统阐述了开关磁阻电机的基本原理与设计特点，并得出了新型磁阻电机的单位出力可以与交流感应电机相媲美甚至还略占优势的结论。1983年英国TASC公司推出了Oulton系列通用SRD调速产品，问世不久便受到了各国电气传动界的广泛重视。从1984年开始，我国许多单位先后开展了SRD研究，在借鉴国外经验的基础上，我国SR电机的研究发展很快。2000年，国内100KW以上的SR电机已应用于煤矿的采煤机，目前已将180KW的SR电机应用于地铁机车的牵引，应形成一些SRD系列商品，最
电机装置磁阻的设置与使用 目前,以横向磁场电机为代表的交流永磁电机的应用日益受到人们的关注。但是这一独特的结构会带来分析方法和制造工艺的复杂性,同时,较高的电枢反应电势导致功率因数偏低,以及损耗和变频器容量增加,这些将限制其实际应用。 为利用横向磁场电机结构的优点并避免永磁电机低功率因数带来的问题,横向磁场开关磁阻电机受到了一定的重视。在开关磁阻电机研究中,可使用与普通交流电机功率因数相似的一个表征输入能量转换为有用输出能量比率的概念,其大小与电机最大、最外位置的磁路饱和情况有关,一般可做到0. 6以上。本文着重分析了横向磁场开关磁阻电机电磁设计与重要参数的计算方法。 1理论分析 1. 1电机结构本文设计的由单相模块构成的外转子横向磁场开关磁阻电机单相结构示意图。 二相样机的A、B相转子模块分布在一根轴线上;而定子模块间则错开半个极距,以便使各绕组在电气上分为180°电角度,构成两相绕组。样机的定、转子铁心结构件以及外壳均由合金铝锭经车、铣加工制成。由于电机气隙较小,相数(轴向模块数)考虑的主要因素为外转子结构的机械强度及高速运行时的可靠性。需要说明的是,电机运行时,三相以下的结构不具备自起动的能量。而实际应用中电机也不一定由二相组成。为便于使用更多的独立电机模块进行串联研究,本样机设计为具有双端轴伸的结构。 1. 2转矩密度方程对于开关磁阻电机,由于运行过程中存在磁路饱和与严重的非线性,电机一个周期内产生的磁阻转矩必须使用Ψ( i,θ)曲线在Ψ- i平面上所围的磁共能面积来表示:T =ΔW /Δθ(1)其中:ΔW =∮iΨ(i,θ)d i;Δθ=θr =2π/n p。
霍尔效应及其应用实验 (FB510A型霍尔效应组合实验仪)（亥姆霍兹线圈、螺线管线圈） 实 验 讲 义 长春禹衡时代光电科技有限公司
实验一 霍尔效应及其应用 置于磁场中的载流体，如果电流方向与磁场垂直，则在垂直于电流和磁场的方向会产生一附加的横向电场，这个现象是霍普金斯大学研究生霍尔于1879年发现的，后被称为霍尔效应。如今霍尔效应不但是测定半导体材料电学参数的主要手段，而且利用该效应制成的霍尔器件已广泛用于非电量的电测量、自动控制和信息处理等方面。在工业生产要求自动检测和控制的今天，作为敏感元件之一的霍尔器件，将有更广泛的应用前景。掌握这一富有实用性的实验，对日后的工作将有益处。 【实验目的】 1．了解霍尔效应实验原理以及有关霍尔器件对材料要求的知识。 2．学习用“对称测量法”消除副效应的影响，测量试样的S H I ~V 和M H I ~V 曲线。 3．确定试样的导电类型。 【实验原理】 1．霍尔效应： 霍尔效应从本质上讲是运动的带电粒子在磁场中受洛仑兹力作用而引起的偏转。当带电粒子（电子或空穴）被约束在固体材料中，这种偏转就导致在垂直电流和磁场方向上产生正负电荷的聚积，从而形成附加的横向电场，即霍尔电场H E 。如图1所示的半导体试样，若在X 方向通以电流S I ，在Z 方向加磁场B ，则在Y 方向即试样A A '- 电极两侧就开始聚集异号电荷而产生相应的附加电场。电场的指向取决于试样的导电类型。对图1（a ）所示的N 型试样，霍尔电场逆Y 方向，（b ）的P 型试样则沿Y 方向。即有 ) (P 0)Y (E )(N 0)Y (E H H 型型?>?< 显然，霍尔电场H E 是阻止载流子继续向侧面偏移，当载流子所受的横向电场力H E e ?
开关磁阻电机驱动系统的运行原理及应用（二） （低轴阻发电机参考资料） 1 引言 开关磁阻电机驱动系统(SDR)具有一些很有特色的优点:电机结构简单、坚固、维护方便甚至免维护，启动及低速时转矩大、电流小;高速恒功率区范围宽、性能好，在宽广转速和功率访问内都具有高输出和高效率而且有很好的容错能力。这使得SR电机系统在家用电器、通用工业、伺服与调速系统、牵引电机、高转速电机、航空航天等领域得到广泛应用。 SR电机是一种机电能量转换装置。根据可逆原理，SR电机和传统电机一样，它既可将电能转换为机械能—电动运行，在这方面的理论趋于成熟;也可将机械能转换为电能—发电运行，其内部的能量转换关系不能简单看成是SR电动机的逆过程。本文将从SR电机电动和发电运行这两个角度阐述SR电机的运行原理。 2 电动运行原理 2.1 转矩产生原理 控制器根据位置检测器检测到的定转子间相对位置信息，结合给定的运行命令(正转或反转)，导通相应的定子相绕组的主开关元件。对应相绕组中有电流流过，产生磁场;磁场总是趋于“磁阻最小”而产生的磁阻性电磁转矩使转子转向“极对极”位置。当转子转到被吸引的转子磁极与定子激磁相相重合(平衡位置)时，电磁转矩消失。此时控制器根据新的位置信息，在定转子即将达到平衡位置时，向功率变换器发出命令，关断当
前相的主开关元件，而导通下一相，则转子又会向下一个平衡位置转动;这样，控制器根据相应的位置信息按一定的控制逻辑连续地导通和关断相应的相绕组的主开关，就可产生连续的同转向的电磁转矩，使转子在一定的转速下连续运行;再根据一定的控制策略控制各相绕组的通、断时刻以及绕组电流的大小，就可使系统在最隹状态下运行。 图1 三相sr电动机剖面图 从上面的分析可见，电流的方向对转矩没有任何影响，电动机的转向与电流方向无关，而仅取决于相绕组的通电顺序。若通电顺序改变，则电机的转向也发生改变。为保证电机能连续地旋转，位置检测器要能及时给出定转子极间相对位置，使控制器能及时和准确地控制定子各相绕组的通断，使srm能产生所要求的转矩和转速，达到预计的性能要求。 2.2 电路分析
霍尔效应及其应用实验(FB510A 型霍尔效应组合实验仪) （亥姆霍兹线圈、螺线管线圈） 实 验 讲 义 长春禹衡时代光电科技有限公司
实验一 霍尔效应及其应用 置于磁场中的载流体，如果电流方向与磁场垂直，则在垂直于电流和磁场的方向会产生一附加的横向电场，这个现象是霍普金斯大学研究生霍尔于1879年发现的，后被称为霍尔效应。如今霍尔效应不但是测定半导体材料电学参数的主要手段，而且利用该效应制成的霍尔器件已广泛用于非电量的电测量、自动控制和信息处理等方面。在工业生产要求自动检测和控制的今天，作为敏感元件之一的霍尔器件，将有更广泛的应用前景。掌握这一富有实用性的实验，对日后的工作将有益处。 【实验目的】 1．了解霍尔效应实验原理以及有关霍尔器件对材料要求的知识。 2．学习用“对称测量法”消除副效应的影响，测量试样的S H I ~V 和M H I ~V 曲线。 3．确定试样的导电类型。 【实验原理】 1．霍尔效应： 霍尔效应从本质上讲是运动的带电粒子在磁场中受洛仑兹力作用而引起的偏转。当带电粒子（电子或空穴）被约束在固体材料中，这种偏转就导致在垂直电流和磁场方向上产生正负电荷的聚积，从而形成附加的横向电场，即霍尔电场H E 。如图1所示的半导体试样，若在X 方向通以电流S I ，在Z 方向加磁场B ，则在Y 方向即试样A A '- 电极两侧就开始聚集异号电荷而产生相应的附加电场。电场的指向取决于试样的导电类型。对图1（a ）所示的N 型试样，霍尔电场逆Y 方向，（b ）的P 型试样则沿Y 方向。即有 ) (P 0)Y (E )(N 0)Y (E H H 型型?>?< 显然，霍尔电场H E 是阻止载流子继续向侧面偏移，当载流子所受的横向电场力H E e ?与洛仑兹力B v e ??相等，样品两侧电荷的积累就达到动态平衡，故有
研胳wZprtf 霍尔效应法测量螺线管磁场实验报告 【实验目的】 1?了解霍尔器件的工作特性。 2?掌握霍尔器件测量磁场的工作原理。 3?用霍尔器件测量长直螺线管的磁场分布。 4.考查一对共轴线圈的磁耦合度。 【实验仪器】 长直螺线管、亥姆霍兹线圈、霍尔效应测磁仪、霍尔传感器等。 【实验原理】 1?霍尔器件测量磁场的原理 图1霍尔效应原理 如图1所示，有—N型半导体材料制成的霍尔传感器，长为L,宽为b,厚为d，其四个侧面各焊有一个电 极1、2、3、4。将其放在如图所示的垂直磁场中，沿3、4两个侧面通以电流I，则电子将沿负I方向以速 ur ir u 度运动，此电子将受到垂直方向磁场B的洛仑兹力F m ev e B作用，造成电子在半导体薄片的1测积累 urn 过量的负电荷，2侧积累过量的正电荷。因此在薄片中产生了由2侧指向1侧的电场E H，该电场对电子ur uuu uir n ir 的作用力F H eE H，与F m ev e B反向，当两种力相平衡时，便出现稳定状态，1、2两侧面将建立起 稳定的电压U H，此种效应为霍尔效应，由此而产生的电压叫霍尔电压U H , 1、2端输出的霍尔电压可由 数显电压表测量并显示出来。 如果半导体中电流I是稳定而均匀的，可以推导出 式中，R H为霍耳系数，通常定义K H R H /d , 由R H和K H的定义可知，对于一给定的霍耳传感器，R H和K H有唯一确定的值，在电流I不变的情况下, U H R H U H满足： 世K H IB , d K H称为灵敏度。
研 島加吋 与B有一一对应关系。 2?误差分析及改进措施 由于系统误差中影响最大的是不等势电势差，下面介绍一种 方法可直接消除不等势电势差的影响，不用多次改变B、丨方 向。如图2所示，将图2中电极2引线处焊上两个电极引线5、6,并在5、6间 连接一可变电阻，其滑动端作为另一引出线2, 将线路完全接通后，可以调节 滑动触头2，使数字电压表所测电压为零，这样就消除了1、2两引线间的不等 势电势差，而且还可以测出不等势电势差的大小。本霍尔效应测磁仪的霍尔电 压测量部分就采用了这种电路，使得整个实验过程变得较为容易操作，不过实 验前要首先进行霍尔输出电压的调零， 以消除霍尔器件的不等位电势”。 在测量过程中，如果操作不当，使霍尔元件与螺线管磁场不垂直，或霍尔元件中电流与磁场不垂直，也会引入系统误差3?载流长直螺线管中的磁场 从电磁学中我们知道，螺线管是绕在圆柱面上的螺旋型线圈。对于密绕的螺线管来说，可以近似地看成是 一系列园线圈并排起来组成的。如果其半径为R、总长度为L,单位长度的匝数为n,并取螺线管的轴线 为x轴，其中心点0为坐标原点，贝U (1)对于无限长螺线管L 或L R的有限长螺线管，其轴线上的磁场是一个均匀磁场，且等于： uu B o o NI 式中0――真空磁导率；N ――单位长度的线圈匝数；I ――线圈的励磁电流。 (2)对于半无限长螺线管的一端或有限长螺线管两端口的磁场为： uu 1 B! —oNI 2 即端口处磁感应强度为中部磁感应强度的一半，两者情况如图3所示。 图2 图3
第二章开关磁阻电机及其调速系统 2.1 开关磁阻电机的发展概况 磁阻式电机诞生于160年前，一直被认为是一种性能不高的电机。然而通过近20年的研究与改进，使磁阻式电机的性能不断提高，目前已能在较大功率范围内不低于其它型式的电机[9]。 70年代初，美国福特电动机（Ford Motor）公司研制出最早的开关磁阻电机调速系统。其结构为轴向气隙电动机、晶闸管功率电路，具有电动机和发电机运行状态和较宽范围调速的能力，特别适用于蓄电池供电的电动车辆的传动。 70年代中期，英国里兹（Leeds）大学和诺丁汉（Nottingham）大学，共同研制以电动车辆为目标的开关磁阻电机调速系统。样机容量从10W至50KW，转速从750 r/min至10000 r/min，其系统效率和电机利用系数等主要指标达到或超过了传统传动系统。该产品的出现，在电气传动界引起了不小的反响。在很多性能指标上达到了出人意料的高水平，整个系统的综合性能价格指标达到或超过了工业中长期广泛使用的一些变速传动系统。 近年来，国内外已有众多高校、研究所和企业投入了开关磁阻电机调速系统的研究、开发和制造工作。至今已推出了不同性能、不同用途的几十个系列的产品，应用于纺织、冶金、机械、汽车等行业中。 目前，在汽车行业意大利FIAT公司研制的电动车和中国第二汽车制造厂研制的电动客车都采用了开关磁阻电机。SRM是没有任何形式的转子线圈和永久磁铁的无刷电动机，它的定子磁极和转子磁极都是凸的。由于SRM具有集中的定子绕组和脉冲电流，其功率变换器可以采用更可靠的电路拓扑形式。SRM具有简单可靠、在较宽转速和转矩范围内高效运行、控制灵活、可四象限运行、响应速度快、成本较低等优点，这是其它调速系统难以比拟的，作为具有潜力的电动车电气驱动系统日益受到重视。然而目前SRM还存在转矩波动大、噪声大、需要位置检测器、系统非线性等缺点，所以，它的广泛应用还受到限制。 2.2 开关磁阻电机的基本结构与特点 开关磁阻电机为定、转子双凸极可变磁阻电机。其定、转子铁心均由硅钢片
霍尔效应测磁场 霍尔效应是导电材料中的电流与磁场相互作用而产生电动势的效应。1879 年美国霍普金斯大学研究生霍尔在研究金属导电机理时发现了这种电磁现象， 故称霍尔效应。后来曾有人利用霍尔效应制成测量磁场的磁传感器，但因金属 的霍尔效应太弱而未能得到实际应用。随着半导体材料和制造工艺的发展，人 们又利用半导体材料制成霍尔元件，由于它的霍尔效应显著而得到实用和发 展，现在广泛用于非电量的测量、电动控制、电磁测量和计算装置方面。在电 流体中的霍尔效应也是目前在研究中的“磁流体发电”的理论基础。近年来，霍尔效应实验不断有新发现。1980年原西德物理学家冯·克利青研究二维电子气系统的输运特性，在低温和强磁场下发现了量子霍尔效应，这是凝聚态物理领域最重要的发现之一。目前对量子霍尔效应正在进行深入研究，并取得了重要应用，例如用于确定电阻的自然基准，可以极为精确地测量光谱精细结构常数等。 在磁场、磁路等磁现象的研究和应用中，霍尔效应及其元件是不可缺少的，利用它观测磁场直观、干扰小、灵敏度高、效果明显。 【实验目的】 1．霍尔效应原理及霍尔元件有关参数的含义和作用 2．测绘霍尔元件的V H—Is，了解霍尔电势差V H与霍尔元件工作电流Is、磁感应强度B之间的关系。 3．学习利用霍尔效应测量磁感应强度B及磁场分布。 4．学习用“对称交换测量法”消除负效应产生的系统误差。 【实验原理】 霍尔效应从本质上讲，是运动的带电粒子在 磁场中受洛仑兹力的作用而引起的偏转。当带电 粒子（电子或空穴）被约束在固体材料中，这种 偏转就导致在垂直电流和磁场的方向上产生正 负电荷在不同侧的聚积，从而形成附加的横向电 场。如图13-1所示，磁场B位于Z的正向，与 之垂直的半导体薄片上沿X正向通以电流Is（称 为工作电流），假设载流子为电子（N型半导体材 料），它沿着与电流Is相反的X负向运动。 由于洛仑兹力f L作用，电子即向图中虚线 箭头所指的位于y轴负方向的B侧偏转，并使B 侧形成电子积累，而相对的A侧形成正电荷积累。 与此同时运动的电子还受到由于两种积累的异种电荷形成的反向电场力f E的作用。随着电荷积累的增加，f E增大，当两力大小相等（方向相反）时，f L=－f E，则电子积累便达到动态平衡。这时在A、B两端面之间建立的电场称为霍尔电场E H，相应的电势差称为霍尔电势V H。 设电子按均一速度v，向图示的X负方向运动，在磁场B作用下，所受洛仑兹力为：
开关磁阻电机的基本学习内容 1 开关磁阻电机的基本原理以及结构 开关磁阻电动机(Switched Reluctance Motor ，简称SRM) 定转子为双凸极结构，铁心均由普通硅钢片叠压而成，其定子极上有集中绕组，径向相对的两个绕组串联构成一相，转子非永磁体，其上也无绕组[1,3]。SRM 的定转子极数必须满足如下约束关系： s r s N =2km N = N + 2k (1-1) 其中，Ns ，Nr 分别为电机定、转子数；m 为电机相数值减1；k 为一常数。以下图1-1所示一个典型四相8/6极SRM 为例，相数为4，因而m=3，取k=1，则Ns=6，Nr=8。m 及k 值越高，越利于高控制性能控制，但相应成本越高，结构越复杂。目前技术较为成熟，发展较为迅速的产品多为三、四相SRM [2]。
图1-1即为一典型四相8/6结构的SRM电机本体及其不对称功率变换器主电路的示意图（图1-1在末尾手画）。为表述清晰，图中仅画出不对称半桥电路的一相，其他各相均与该相相同，并省略了相应的驱动及检测电路。完整的开关磁阻电机调速系统（Switched Reluctance Motor Drive，简称SRD）则由SRM、功率变换器、控制器、位置检测器等四大部分组成，如下图1-2示。 SRM可以认为是同步电机的一个分支，它运行时遵循磁阻最小原理，同步进电机较为类似[2,30]。其具体运行原理如下：首先要保证励磁相的定子凸极和最近的转子凹极中心线不重合，也即初始位移不能位于磁阻最小位置。通以交流电后，经过一个整流桥变为直流电源，当开关S1和S2开通时，AA’相通电励磁，产生一个磁拉力。在该电磁力的轴向分量作用下，产生电磁转矩，凸极转子铁心趋向于旋转到定转子极轴线B-B’与A-A’重合的位置；而电磁力的径向力分量则造成定子的“变形”，这也是产生转矩脉动和电机噪声的根本原因之一。在该过程中电机吸收电能。关断S1和S2，开通BB’相，此时AA’相经续流二极管VD1、VD2将电能回馈给电源，同时BB’相趋向运行到定转子极轴线C-C’与B-B’重合的位置。以此类推，顺次给A→B→C→D相循环励磁，在惯性和轴向力的作用下，转子将一直逆着励磁顺序旋转，从而完成自同步运行。同理若改变励磁顺序为C→B→A→D，则转子沿顺时针方向转动。由此可以看出， SRM与直流电机不同，其运行方向与相电流方向无关，而仅与相绕组通电顺序有关。 图1-2开关磁阻电机调速系统构成
开关磁阻电机调速系统 开关磁阻电机调速系统(Switched Reluctance Driver，简称SRD)是以现代电力电子与微机控制技术为基础的机电—体化产品。除了具有显著的节能效果外，开关磁阻电机的理论研究和实践证明，它与常用的三相异步电动机相比还有以下优点: 1.电机结构简单、坚固，制造工艺简单，成本低，可工作于极高转速；定子线圈嵌放容易，端部短而牢固，工作可靠，能适用于各种恶劣、高温甚至强振动环境； 2.起动转矩大，低速性能好，无感应电动机在起动时所出现的冲击电流现象； 3.调速范围宽，控制灵活，易于实现各种特殊要求的转矩；λ λ 4.在宽广的转速和功率范围内都具有高效率； 5.损耗主要产生在定子，电机易于冷却，转子无永磁体，无高温退磁现象；λλ 6.转矩方向与电流方向无关，从而可最大限度简化功率变换器，降低系统成本； 7.功率变换器不会出现直通故障，可靠性高；λ λ 8.能四象限运作，具有较强的再生制动能力； 开关磁阻电机调速系统(SRD) 开关磁阻电机调速系统(SRD)是当今世界最新、性能价格比最高的调速系统。它是一种基于改变供电电源频率的调速方式——交流变频调速系统应运而生。而开关磁阻电机调速系统(又称开关磁阻电机驱动系统)简称SRD系统,是它们中崭新的一种系统,并且已经是智能化和模块化,不仅调速性优越，而且各种保护功能也很完善，已在很多方面大量使用。这项技术一经问世，便以其宽广的调速范围，良好的机械特性，卓越的启动制动性能，节能，易维护等一系列突出优点而引起电气及其他行业的关注。SRD系统是磁阻电动机和电力电子技术相结合而产生的一种机电一体化装置,主要由SRM开关磁阻电动机、功率变换器、单片机(或DSP 芯片)、电流及位置检测器等五大部分组成。其组成与特点： 1.1开关磁阻电动机(Switched Reluctance Motor,简称SRM) 是系统中实现能量转换的部件, 它与传统的磁阻电动机相比，具有本质的区别。在结构上，SRM采用双凸极形式，即定子、转子均为凸极式构造；定子线圈采用集中式而不是分布式绕组；加在定子绕组上的电压为不连续的矩形波而非连续的正弦波。转子仅由硅钢片叠压而成，既无绕组也无永磁体，定子各极上绕有集中绕组。图2所示为8/6极（定子八极、转子六极）四相SRM剖面图. SRM有两种独特的运行方式：低速时采用电流斩波方式；高速时采用单脉冲角度控制方式。在电流斩波方式中，系统是通过调节相绕组电流的大小来控制转矩，因此能准确知道绕组中实际电流的大小，对电流进行反馈是很必要的；在角度位置控制方式中，系统通过调节触发角和关断角来实现对转矩的控制，此时电流己不再作为控制量，但为了防止系统过载或故障则要进行过流保护，所以系统中需要进行电流检测。 1.11开关磁阻电动机(SRM)工作原理遵循“磁阻最小原理”，通电后，磁路有向磁阻最小路径变或化的趋势。当转子凸极与电子凸极错位时，气隙大、磁阻大：一旦定子磁极绕组通
【最新整理，下载后即可编辑】 实 验 报 告 学生姓名： 学 号： 指导教师： 实验地点： 实验时间： 一、实验室名称：霍尔效应实验室 二、 实验项目名称：霍尔效应法测磁场 三、实验学时： 四、实验原理： （一）霍耳效应现象 将一块半导体（或金属）薄片放在磁感应强度为B 的磁 场中，并让薄片平面与磁场方向（如Y 方向）垂直。如在薄片的横向（X 方向）加一电流强度为H I 的电流，那么在与磁场方向和电流方向垂直的Z 方向将产生一电动势H U 。 如图1所示，这种现象称为霍耳效应，H U 称为霍耳电压。霍耳发现，霍耳电压H U 与电流强度H I 和磁感应强度B 成正比，与磁场方向薄片的厚度d 反比，即 d B I R U H H = （1） 式中，比例系数R 称为霍耳系数，对同一材料R 为一常数。因成品霍耳元件（根据霍耳效应制成的器件）的d 也是一常数，故d R /常用另一常数K 来表示，有 B KI U H H = （2） 式中，K 称为霍耳元件的灵敏度，它是一个重要参数，表示该元件在单位磁感应强度和单位电流作用下霍耳电压的大小。如果霍
耳元件的灵敏度K 知道（一般由实验室给出），再测出电流H I 和霍耳电压H U ，就可根据式 H H KI U B = （3） 算出磁感应强度B 。 图 1 霍 耳 效 应 示 意 图 图2 霍耳效应解释 （二）霍耳效应的解释 现研究一个长度为l 、宽度为b 、厚度为d 的N 型半导体制成的霍耳元件。当沿X 方向通以电流H I 后，载流子（对N 型半导体是电子）e 将以平均速度v 沿与电流方向相反的方向运动，在磁感应强度为B 的磁场中，电子将受到洛仑兹力的作用，其大小为 evB f B = 方向沿Z 方向。在B f 的作用下，电荷将在元件沿Z 方向的两端面堆积形成电场H E （见图2），它会对载流子产生一静电力E f ，其大小为 H E eE f = 方向与洛仑兹力B f 相反，即它是阻止电荷继续堆积的。当B f 和E f 达到静态平衡后，有E B f f =，即b eU eE evB H H /==，于是电荷堆积的两端面（Z 方向）的电势差为 vbB U H = （4）
第12讲磁场三难之霍尔效应 题一：利用如图所示的方法可以测得金属导体中单位体积内的自由电子数n，现测得一块横截面为矩形的金属导体的宽为b，厚为d，并加有与侧面垂直的匀强磁场B，当通以图示方向的电流I时，用电压表测得导体上、下表面间的电压为U。已知自由电子的电荷量为e，则该导体单位体积内的自由电子数为________。 C为其四个侧面，如图所示，已知题二：一导体材料样品的体积为a×b×c，'A、C、A、' 导体样品中载流子是自由电子，且单位体积中的自由电子数为n，电阻率为ρ，电子的电荷量为e，沿x方向通有电流I。 （1）导体A'、A两个侧面之间的电压大小为________，导体中自由电子定向移动的速率是________； （2）将该导体样品放在匀强磁场中，磁场方向沿z轴正方向，则导体样品侧面C的电势 C的电势； ________(填“高于”“低于”或“等于”)侧面' C两侧面间的电势差（3）在（2）中，达到稳定状态时，沿x方向电流仍为I，若测得C、' 为U，则匀强磁场的磁感应强度为________。 题三：如图所示，a、b、c分别表示长方体导体板的长、宽、高，将导体板置于垂直导体板前后两个表面向外、磁感应强度大小为B的匀强磁场中，当电流由左侧面向右侧面通过导体板时，将在导体板的上、下两表面之间产生霍尔电压U H，下列说法中正确的是（） A．导体板下表面电势高于其上表面电势 B．若保持通过导体板的电流恒定不变，只将导体板的高度c减半，则U H将减半 C．若保持导体板左、右两端所加电压恒定不变，只将导体板的宽度b减半，则U H将不变D．若保持导体板左、右两端所加电压恒定不变，只将导体板的长度a减半，则U H将变为原来的2倍 题四：利用霍尔效应制作的元件，广泛应用于测量和自动控制等领域。如图是霍尔元件的工作原理示意图，磁感应强度为B的匀强磁场垂直于霍尔元件的工作面向下，通入图示方向的电流I，C、D两侧就会形成电势差U CD，下列说法中正确的是（）
开关磁阻电机 Switched Reluctance Drive system, SRD 开关磁阻电机驱动系统(Switched Reluctance Drive system, SRD)具有一些很有特色的优点:电机结构简单、坚固、维护方便甚至免维护，起动及低速时转矩大、电流小;高速恒功率区范围宽、性能好，在宽广转速和功率范围内都具有高输出和高效率而且有很好的容错能力。这使得SR电机驱动系统在家用电器、通用工业、伺服与调速系统、牵引电机、高转速电机、航空航天等领域得到广泛应用。 SR电机是一种机电能量转换装臵。根据可逆原理，SR电机和传统电机一样，它既可将电能转换为机械能——电动运行，在这方面的理论趋于成熟;也可将机械能转换为电能——发电运行，其内部的能量转换关系不能简单看成是SR 电动机的逆过程。 开关磁阻电机的发展概况和发展趋势 “开关磁阻电机(Switched reluctance motor)”一词源见于美国学者S.A.Nasarl969年所撰论文，它描述了这种电机的两个基本特征：①开关性——电机必须工作在一种连续的开关模式，这是为什么在各种新型功率半导体器件可以获得后这种电机才得以发展的主要原因；②磁阻性——它是真正的磁阻电机，定、转子具有可变磁阻磁路，更确切地说，是一种双凸极电机。开关磁阻电机的概念实际非常久远，可以追溯到19世纪称为“电磁发动机”的发明，这也是现代步进电机的先驱。在美国，这种电机常常被称为“可变磁阻电机(variable reluctance motor, VR电机)”一词, 但是VR电机也是步进电机的一种形式，容易引起混淆。有时人们也用“无刷磁阻电机(Brushless reluctance motor)”一词，以强调这种电机的无刷性。“电子换向磁阻电机(Electronically commutated reluctance motor)”一词也曾采用，从工作原理来看，甚至比“开关磁阻”的说法更准确—些，但也容易与电子换向的水磁直流电机相混淆。毫无疑问，正是由于英国P．J．Lawrenson教授及其同事们的杰出贡献，赋予了现代SR电机新的意义，开关磁阻电机一词也因此逐渐为人们所接受和采用。 从电机结构和运行原理上看，SR电机与大步距角的反应式步进电机十分相似，因此有人将SR电机看成是一种高速大步距角的步进电机。但事实上，两者是有本质差别的，这种差别体现在电机设计、控制方法、性能特性和应用场合等方面，见表11-1。
开关磁阻电动机原理 资料来源：https://www.doczj.com/doc/973464003.html,/zindex01.html 开关磁阻电动机（SR）是近些年发展的新型调速电机，结构简单结实、调速范围宽且性能好，现已广泛用在仪器仪表、家电、电动汽车等领域。 下面通过一个开关磁阻电动机原理模型来介绍工作原理。 双凸极结构 磁阻电机的定子铁芯有六个齿极，由导磁良好的硅钢片冲制后叠成，见下图。 磁阻电机定子铁芯 磁阻电机的转子铁芯有四个齿极，由导磁良好的硅钢片冲制后叠成，见下图。 磁阻电机转子铁芯
与普通电机一样，转子与定子直接有很小缝隙，转子可在定子内自由转动，见下图。 双凸极结构的定子铁芯与转子铁芯 由于定子与转子都有凸起的齿极，这种形式也称为双凸极结构。在定子齿极上绕有线圈（定子绕组），是向电机提供工作磁场的励磁绕组。 定子铁芯上有励磁绕组 在转子上没有线圈，这是磁阻电机的主要特点。在讲电动机工作原理时常用通电导线在磁场中受力来解释电动机旋转的道理，但磁阻电机转子上没有线圈，也无“鼠笼”，那是靠什么力推动转子转动呢？磁阻电动机则是利用磁阻最小原理，也就是磁通总是沿磁阻最小的路径闭合，利用齿极间的吸引力拉动转子旋转。
三相6/4结构工作原理 下面通过图示来说明转子的工作原理，下面是磁阻电动机的正视图，定子六个齿极上绕有线圈，径向相对的两个线圈连接在一起（标有紫色圆点的线端连接在一起），组成一“相”，该电机有3相，结合定子与转子的极数就称该电机为三相6/4结构。在下图标注的A相、B相、C相线圈仅为后面分析磁路带来方便，并不是连接普通的三相交流电。 磁阻电机励磁绕组分布图 在下面有一组磁阻电动机运转原理动画的截图，从中我们将看到磁阻电动机是如何转动起来的。A相、B相、C相线圈由开关控制电流通断，图中红色的线圈是通电线圈，黄色的线圈没有电流通过；通过定子与转子的深蓝色线是磁力线；约定转子启动前的转角为0度。 从左面图起，A相线圈接通电源产生磁通，磁力线从最近的转子齿极通过转子铁芯，磁力线可看成极有弹力的线，在磁力的牵引下转子开始异时针转动；中间图是转子转了10度的图，右面图是转到20度的图，磁力一直牵引转子转到30度为止，到了30度转子不再转动，此时磁路最短。 磁阻电机工作原理示意图-1 为了使转子继续转动，在转子转到30度前已切断A相电源在30度时接通B相电源，磁通从最近的转子齿极通过转子铁芯，见下左图，于是转子继续转动。中间图是转子转到40度的图，右面图是转到50度的图，磁力一直牵引转子转到60度为止。 磁阻电机工作原理示意图-2
霍尔效应及磁场的测定 近年来，在科研和生产实践中，霍尔传感器被广泛应用于磁场的测量，它的测量灵敏度高，体积小，易于在磁场中移动和定位。本实验利用霍尔传感器测量通电螺线管内直流电流与霍尔传感器输出电压之间的关系，证明霍尔电势差与螺线管内的磁感应强度成正比，从而掌握霍尔效应的物理规律；用通电螺线管中心点磁场强度的理论计算值作为标准值来校准霍尔元件的灵敏度；用霍尔元件测螺线管内部的磁场沿轴线的分布。 【实验目的与要求】 1.了解霍尔传感器的工作原理，学习测定霍尔传感器灵敏度的方法； 2.掌握用霍尔传感器测量螺线管内磁感应强度沿轴线方向的分布。 【实验原理】 一、霍尔效应 图8-1 霍尔效应原理图 把矩形的金属或半导体薄片放在磁感应强度为B 的磁场中，薄片平面垂直于磁场方向。如图8-1所示，在横向方向通以电流I ，那么就会在纵向方向的两端面间出现电位差，这种现象称为霍尔效应，两端的电压差称为霍尔电压，其正负性取决于载流子的类型。(图8-1载流子为带负电的电子，是N 型半导体或金属)，这一金属或半导体薄片称为霍尔元件。假设霍尔元件由N 型半导体制成，当霍尔元件上通有电流时，自由电子运动的方向与电流I 的流向相反的。由于洛伦兹力B v e F m ?-=的作用，电子向一侧偏转，在半导体薄片的横 向两端面间形成电场，称为霍尔电场H E ，对应的电势差称为霍尔电压U H 。电子在霍尔电场H E 中所受的电场力为H H E e F -=，当电场力与磁场力达到平衡时，有 ()()0=?-+-B v e E e H B v E H ?-=
若只考虑大小，不考虑方向有 E H =vB 因此霍尔电压 U H =wE H =wvB (1) 根据经典电子理论，霍尔元件上的电流I 与载流子运动的速度v 之间的关系为 I=nevwd (2) 式中n 为单位体积中的自由电子数，w 为霍尔元件纵向宽度，d 为霍尔元件的厚度。由式(1)和式(2)可得 IB K IB d R end IB U H H H =?? ? ??== (3) 即 I K U B H H = (4) 式中en R H 1=是由半导体本身电子迁移率决定的物理常数，称为霍尔系数，而K H 称为霍尔 元件的灵敏度。在半导体中，电荷密度比金属中低得很多，因而半导体的灵敏度比金属导体大得多，所以半导体中，电荷密度比金属中低得多，因而半导体的灵敏度比金属导体大得多，所以半导体能产生很强的霍尔效应。对于一定的霍尔元件，K H 是一常数，可用实验方法测定。 图8-2 SS95A 型集成霍尔传感器结构图 虽然从理论上讲霍尔元件在无磁场作用(B =0)时，U H =0，但是实际情况用数字电压表测量并不为零，这是由于半导体材料结晶不均匀、各电极不对称等引起附加电势差，该电势差U HO 称为剩余电压。随着科技的发展，新的集成化(IC)器件不断被研制成功，本实验采用SS95A 型集成霍尔传感器(结构示意图如图8-2所示)是一种高灵敏度传感器，它由霍尔元件、放大器和薄膜电阻剩余电压补偿器组成。其特点是输出信号大，并且已消除剩余电压的影响。SS95A 型集成霍尔传感器有三根引线，分别是：“V +”、“V －”、“V out ”。其中“V +”和“V －”构成“电流输入端”，“V out ”和“V －”构成“电压输出端”。由于SS95A 型集成霍尔传感器它的工作电流已设定，被称为标准工作电流，使用传感器时，必须使工作电流处于该标准状态。在实验时，只要在磁感应强度为零(B =0)条件下，“V out ”和“V －”之间的电压为2.500V ，则传感器就处于标准工作状态之下。
2010 年5 月 摘要 开关型磁阻电动机驱动系统(Switched Reluctance Drive，简称SRD电动机)。是20世纪80年代迅猛发展起来的一种新型调速电机驱动系统。它是由功率变换电路、双凸极磁阻电机、控制器及位置检测器构成。它的结构极其简单，调速范围宽，调速性能优异，而且在整个调速范围内都具有较高的效率，系统可靠性高，是各国研究和开发的热点之一。 本文介绍了开关磁阻电机的发展历史,应用领域以及它的优点；对三相6/4结构的开关磁阻电机与四相8/6结构的开关磁阻电机进行了比较；对开关磁阻电机的电磁设计与参数优化进行了分析与研究，简单介绍了ANSYS软件在开关磁阻电机电磁分析中的应用；提出8/6结构开关磁阻电机的一种设计方案；并对开关磁阻电机的磁通波形和电机损耗进行了分析。 关键词: 开关磁阻电机，磁场，电磁设计，参数优化
ABSTRACT The switched reluctance drive (SRD) is a new-type drived-electromotor system which develops rapidly since 1980, and consists of power converter circuits、the doubly-salient reluctance motor、the controller and the examination of position. The structure of the SRD is simple. It has a wide range and excellent performance in speed. It also has a high efficiency and high reliability. So the SRD is one of the hot spots which is studied and designed all over the world. This thesie introduced the SRD development history, the application domain as well as its merit; comparison to the three-phase 6/4 structure SRD with four-phase 8/6 structure SRD overall performance. also analysis and research SRD electromagnetism design and parameter optimization, and introduced ANSYS software in SRD electromagnetism analysis application; Proposes 8/6 structure SRD one kind of design proposal; And analysis to the switched reluctance drive magnetic flux profile and the loss of machine. Keywords:switched reluctance motor, magnetic field, electromagn- etism design, parameter optimization